Phase correcting device, distance measuring device, phase fluctuation detecting device and phase correction method

ABSTRACT

A phase correcting device includes a local oscillator that includes an all digital phase-locked loop configured to output a local oscillation signal, a first phase detector configured to detect a phase of the local oscillation signal to output the phase of the local oscillation signal, a reference phase device configured to generate a quasi-reference phase corresponding to a reference phase of the local oscillation signal to output the quasi-reference phase, based on a reference clock, a second phase detector configured to detect a fluctuation amount of a phase of the local oscillator, based on the phase detected by the first phase detector and the quasi-reference phase, and a correction circuit configured to correct the phase of the inputted signal by using a detection result of the second phase detector.

CROSS REFERENCE TO RELATED APPLICATION

This application is based upon and claims the benefit of priority fromthe prior Japanese Patent Application No. 2020-049734 filed in Japan onMar. 19, 2020; the entire contents of which are incorporated herein byreference.

FIELD

An embodiment described herein relates generally to a phase correctingdevice, a distance measuring device, a phase fluctuation detectingdevice and a phase correction method.

BACKGROUND

In recent years, keyless entry systems that make it easy to lock andunlock cars have been adopted by many automobiles. According to thetechnique, a user of an automobile can lock and unlock doors by usingcommunication between a key of the automobile and the automobile.Further, in recent years, a smart key system that allows a user to lockand unlock a door or start an engine without touching a key has alsobeen widely used.

However, there have been many cases where an attacker who carries out aso-called relay attack invades the communication between a key and anautomobile, and steals a vehicle or articles in the vehicle. Therefore,as a defense measure against the aforementioned attack (so-called relayattack), a measure for measuring the distance between the key and theautomobile, and prohibiting the control of the vehicle by communicationwhen the distance is determined to be a predetermined distance or moreis being studied.

There are a time detection method, a frequency difference detectionmethod, a phase detection method and the like, as distance measurementmethods, but due to the ease of implementation, a distance measuringsystem is receiving attention which employs a communication type phasedetection method that obtains the distance between respective devices bycommunication between the respective devices. However, since referencesignals between the respective devices independently operate, theinitial phases differ from each other, and therefore distancemeasurement accuracy is generally greatly deteriorated in thecommunication type phase detection method. Therefore, there is proposedthe technique that enables distance measurement by transmitting phaseinformation detected in one device to the other device. According to theproposal, it is possible to calculate a highly accurate distance byperforming a predetermined operation by using phase information of thesignals detected by receiving units of two distance measuring devicesthat form a pair.

Note that in the proposal, accurate distance measurement is enabled onthe precondition that the initial phase does not fluctuate in the localoscillator in the distance measuring device.

Since the distance measuring device is also mounted on a key side, thereis a demand for extending the battery life of the key, and low powerconsumption of the distance measuring device is required. Since most ofthe power consumption of the distance measuring device is consumed bywireless units, reduction in power consumption of the wireless units isrequired. The power consumption of the wireless units strongly dependson the architecture of the wireless units. A configuration using adigital-controlled oscillator (DCO) direct modulation method(hereinafter, also referred to as a DCO direct modulation method) for atransmission unit, and a super-heterodyne (SH) method (hereinafter, alsoreferred to as an SH method) or a Low-IF reception method for areception unit is widely known as a configuration of low powerconsumption. Therefore, it is desired to realize a distance measuringdevice by the configuration using a DCO direct modulation method for thetransmission unit, and using an SH method for the reception unit.

However, when distance measurement is performed by using a DCO directmodulation method for the transmission unit, and using an SH method forthe reception unit, the initial phase fluctuates in the local oscillatorin the distance measuring device. Therefore, accurate distancemeasurement cannot be performed with the distance measuring device usinga DCO direct modulation method for the transmission unit, and using anSH method for the reception unit.

Note that the fluctuation of the initial phase in the local oscillatormay have an adverse effect on not only the distance measuring device butalso various devices that detect the phases of the signals inputted byusing the local oscillator.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram illustrating a distance measuring systemconfigured by distance measuring devices each including a phasefluctuation detecting device and a phase correcting device according toan embodiment;

FIG. 2 is a configurational diagram of a wireless circuit of a distancemeasuring system in a related art that carries out distance measurementbetween two devices by a communication type phase detection method;

FIG. 3 is an explanatory diagram illustrating an “8 alternations”distance measuring sequence in a case where the distance measuringdevices each using a DCO direct modulation method for a transmissionunit, and a heterodyne method for a reception unit;

FIG. 4 is a graph illustrating changes of phases of signals S2 and S5 inthe “8 alternations” distance measuring sequence, by plotting a time ina horizontal axis, and plotting a phase θ in a vertical axis;

FIG. 5 is an explanatory diagram illustrating settings of a device 1Aand a device 2A between a time t₁ and a time t₂ of FIG. 3;

FIG. 6 is a diagram for explaining operations in the devices of FIG. 2;

FIG. 7 is a diagram in which arrows explaining information on five kindsof phase differences are added to a graph similar to the graph of FIG.4;

FIG. 8 is a diagram in which arrows explaining information on five kindsof phase differences are added to a graph similar to the graph of FIG.4;

FIG. 9 is a block diagram illustrating the distance measuring deviceincluding the phase fluctuation detecting device and the phasecorrecting device according to the embodiment;

FIG. 10 is a circuit diagram illustrating a more specific configurationof part of mpl2 and a phase calculator phscalc2;

FIG. 11 is a block diagram illustrating one example of a specificconfiguration of a timing generation circuit 40;

FIG. 12 is a graph by a similar description method to a descriptionmethod of the graph in FIG. 7;

FIG. 13 is a similar graph to the graph in FIG. 12;

FIG. 14 is an explanatory diagram for explaining a difference between aphase ϕ_(tx2) of mpl20 and a quasi-reference phase ϕ₀ obtained from anoutput of a reference phase device mpl00;

FIG. 15 is an explanatory diagram illustrating a waveform of a phasedifference in FIG. 14 by adding a waveform of a phase difference towhich equation (58) is applied to the waveform of the phase differencein FIG. 14 by a dash-dotted line; and

FIG. 16 is an explanatory diagram similar to the explanatory diagram inFIG. 14.

DETAILED DESCRIPTION

A phase correcting device of an embodiment includes a local oscillatorthat includes an all digital phase-locked loop configured to generate alocal oscillation signal based on a reference clock, and is configuredto give the local oscillation signal to a device configured to detect aphase of an inputted signal, a first phase detector included in the alldigital phase-locked loop, and configured to detect a phase of the localoscillation signal to output the phase of the local oscillation signal,a reference phase device configured to generate a quasi-reference phasecorresponding to a reference phase of the local oscillation signal at atime of an initial setting of the local oscillator to output thequasi-reference phase, based on the reference clock, a second phasedetector configured to detect a fluctuation amount of a phase of thelocal oscillator, based on the phase detected by the first phasedetector and the quasi-reference phase, and a correction circuitconfigured to correct the phase of the inputted signal by using adetection result of the second phase detector.

Hereinafter, the embodiment of the present invention will be describedwith reference to the drawings.

Embodiment

FIG. 1 is a block diagram illustrating a distance measuring systemconfigured by distance measuring devices each including a phasefluctuation detecting device and a phase correcting device according tothe embodiment. Distance measurement obtaining a distance between adevice 1 and a device 2 is enabled by performing transmission andreception of a single wave signal between the devices 1 and 2 in FIG. 1.Note that the single wave signal is a signal of a single frequency suchas an unmodulated carrier.

In the present embodiment, an example in which the phase fluctuationdetecting device and the phase correcting device are applied to thedistance measuring device is explained, but it is also possible to applythe phase fluctuation detecting device and the phase correcting deviceto various devices that detect phases of inputted signals other than thedistance measuring device. For example, application to a positioningdevice is also possible.

FIG. 2 is a configurational diagram of a wireless circuit of a distancemeasuring system in a related art that performs distance measurementbetween two devices by a communication type phase detection method. InFIG. 1 and FIG. 2, same components are assigned with same referencesigns, and redundant explanation will be omitted for the samecomponents.

First, with reference to FIG. 2, a reason why accurate measurementcannot be performed even when phase information of signals detected byreception units of two distance measuring devices that form a pair isused with distance measuring devices each of a related art using adigital-controlled oscillator (DCO) direct modulation method for atransmission unit, and using a super heterodyne (SH) method for areception unit will be described. Further, FIG. 3 to FIG. 6 are diagramsfor explaining operations in the devices in FIG. 2.

A distance measuring system 100A includes a device 1A and a device 2A.At least one of the device 1A and the device 2A is movable. In thedistance measuring system 100A, a distance between the device 1A and thedevice 2A is calculated based on carrier phase detection. A case whereone of the device 1A and the device 2A calculates the distance based onphase information acquired by the device 1A and the device 2A will beconsidered.

The device 1A transmits a first distance measurement signal (single wavesignal), and the device 2A transmits a second distance measurementsignal (single wave signal). The first and the second distancemeasurement signals reach the device 2A and the device 1A respectivelyvia a propagation path PDLY between the device 1A and the device 2A. Thedevice 1A and the device 2A each include a wireless circuit using a DCOdirect modulation method of low power consumption for a transmissionunit, and using an SH method of low power consumption for a receptionunit.

FIG. 2 illustrates a configuration of simplified wireless units of thedevice 1A and the device 2A. The device 1A includes an oscillator (OSC1)peculiar to the device, a frequency multiplier (mpl1A), an RF frequencyconverter (RFMIX1), a frequency divider (div1), and an intermediate (IF)frequency converter (IFMIX1). The device 2A also includes a wirelessarchitecture similar to the device 1A, and includes an oscillator (OSC2)peculiar to the device, a frequency multiplier (mpl2A), an RF frequencyconverter (RFMIX2), a frequency divider (div2), and an intermediate (IF)frequency converter (IFMIX2). Note that in the devices 1A and 2A, outputsignals of mpl1A and mpl2A become transmission signals of the devices 1Aand 2A, and are also used as local signals (LO signals). In other words,mpl1A and mpl2A respectively configure local oscillators.

Hereinafter, in order to clarify a problem, the device 1A and the device2A are assumed to set transmission frequencies first of all. In otherwords, in an initial setting, for example, the transmission frequenciesof the devices 1A and 2A are respectively set at frequencies obtained bymultiplying the oscillation frequencies of OSC1 and OSC2 by apredetermined multiple k_(L).

An output signal (oscillation signal) S1 (=lo_(x1)) of OSC1 of thedevice 1A can be expressed by equation (1) as follows with a frequencyof an oscillation signal of OSC1 set as f_(x1) and an initial phase setas θ_(x1).lo _(x1)=sin(2πf _(x1) t+θ_(x1))  (1)

The oscillation frequency of OSC1 is multiplied by k_(L) by mpl1A. Aphase ϕ_(tx1) of an output signal S2 of mpl1A is expressed asϕ_(tx1)=2πk _(L) f _(x1) t+θ _(Lx1)  (2).Here, θ_(Lx1) is an initial phase of the output signal S2 of mpl1A. Anoutput of mpl1A is generally generated by a digitally controlledoscillator (DCO) technique and a digital frequency/phase synchronizationtechnique. Note that in mpl1A using a TDC (time to digital converter)for a phase synchronizing unit, θ_(Lx1)=k_(L)θ_(x1) is not generallyestablished. Therefore, in equation (2) described above, an initialphase of the output S2 of mpl1A is defined as θ_(Lx1).

For the device 2A, a similar transmission frequency setting is alsoperformed. An output signal S4 (=lo_(x2)) of OSC2 of the device 2A canbe expressed by equation (3) as follows with a frequency of anoscillation signal of OSC2 set as f_(x2), and an initial phase set asθ_(x2).lo _(x2)=sin(2πf _(x2) t+θ _(x2))  (3)

In mpl2A, the oscillation frequency of OSC2 is multiplied by k_(L). Aphase ϕ_(tx2) of an output signal S5 of mpl2A is expressed asϕ_(tx2)=2πk _(L) f _(x2) t+θ _(Lx2)  (4).Here, θ_(Lx2) is an initial phase of the output of mpl2A. For the outputof mpl2A, θ_(Lx2)=k_(L)θ_(x2) is not generally established, either, asin the output of mpl1A. Therefore, in equation (4) described above, theinitial phase of the output of mpl2A is defined as θ_(Lx2).

Patent Literature 1 discloses that in the case of a system of TDD (timedivision duplex) that does not simultaneously carry out transmission andreception, correct distance measurement can be performed by performingexchange of single wave signals between the device 1A and the device 2A.Note that the devices in Patent Literature 1 differ from the devices inFIG. 2 in configuration of the wireless units. Patent Literature 1 showsthat distance measurement can be correctly performed by adopting adistance measurement sequence of “8 alternations” that repeatstransmission and reception of four times each in total between thedevices 1A and 2A by each using two single-wave signals. Transmittingand receiving while changing the frequency like this is enabled bychanging settings of mpl1A and mpl2A in the devices 1A and 2A.

FIG. 3 illustrates an example of an “8 alternations” distancemeasurement sequence in a case where the distance measuring devices eachusing a DCO direct modulation method for the transmission unit, and aheterodyne method for the reception unit are employed. Explaining anorder of alternations by paying attention to transmission signalsregarding the distance measurement sequence in FIG. 3, the order is asfollows. The device 1A transmits signals of a frequency k_(L)f_(x1) attimes t=t₁, and t=t₃, and transmits signals of frequency k_(H)f_(x1) attimes D+t₁, and D+t₃. The device 2A transmits a signal of a frequencyk_(L)f_(x2) twice from a time t₂, and transmits a signal of frequencyk_(H)f_(x2) twice from a time D+t₂.

The device 1A and the device 2A perform transmission after thefrequencies of the transmission signals are respectively set atk_(L)f_(x1) and the frequency of k_(L)f_(x2) (hereinafter, thesefrequencies are also referred to as low frequencies) in the initialsetting. When only transmission of the devices 1A and 2A is considered,a single wave signal of the frequency k_(L)f_(x1) is transmitted fromthe device 1A to the device 2A first, and the device 2A receives thesingle wave signal of the frequency k_(L)f_(x1) from the device 1A. FIG.3 shows that transmission is performed at the time t₁ after it takes apredetermined time period for the device 1A to be set to transmit thesingle wave signal of the frequency k_(L)f_(x1) to the device 2A.

Next, after it takes a predetermined time period for the device 2A to beset to transmit the single wave signal of the frequency k_(L)f_(x2) tothe device 1A, transmission of the single wave signal is performed twiceat the time t₂. Furthermore, a single wave signal of the frequencyk_(L)f_(x1) is transmitted from the device 1A to the device 2A again,and the device 2A receives the single wave signal of the frequencyk_(L)f_(x1) from the device 1A. The device 1 takes a predetermined timeperiod for the transmission, and performs transmission at the time t₃.The signal exchanges end at a time t₄.

FIG. 4 is a graph illustrating changes in phases of the signals S2 andS5 in the “8 alternations” distance measurement sequence, with a timeplotted in a horizontal axis and a phase θ plotted in a vertical axis.Note that numbers shown in parentheses in FIG. 4 correspond to numbersof equations in the description. In the initial phases shown in thevertical axis in FIG. 4, L included in subscripts indicates that thesingle wave signal has a low frequency, x1 indicates the initial phaseof the signal S2, x2 indicates the initial phase of the signal S5, mindicates a case of multiplying the frequency by (k_(L)+m), and numbersin subscript parentheses of the phases θ in FIG. 4 correspond to ordersof a change in frequency from the frequency of the initial settingwithout parentheses. In the description, the same subscripts will beused hereinafter, and H of a subscript in each of signs indicating theinitial phases of the signals S2 and S5 indicates that the single wavesignal has a high frequency.

A dashed straight line (2) in FIG. 4 represents a phase ϕ_(tx1) of theoutput signal S2 of mpl1A of the device 1A, and a dashed straight line(4) represents a phase ϕ_(tx2) of the output signal S5 of a mpl2A of thedevice 2A. The phase ϕ_(tx1) has a linear characteristic having aninclination of 2πK_(L)f_(x1) with the initial phase as θL_(x1). Thephase ϕ_(tx2) has a linear characteristic having an inclination of2πK_(L)f_(x2) with the initial phase as θ_(Lx2).

However, in the distance measuring device in FIG. 2 using a DCO directmodulation method and a heterodyne method, it is necessary to change thefrequencies of the transmission signals of mpl1A and mpl2A at atransmission time and a reception time.

FIG. 5 is an explanatory diagram illustrating settings of the device 1Aand the device 2A between the time t₁ and the time t₂ in FIG. 3. Notethat in a period between the times t₁ and t₂, a reception operation ofthe device 1A is not performed, and therefore unnecessary units foroperation are shown by dash-dotted lines.

In the heterodyne method, a reception signal is converted into an IFfrequency. In an example of FIG. 5, RFMIX2 of the device 2A needs toconvert a reception signal into an IF frequency of approximately−mf_(x2). For this reason, in the device 2A that receives a single wavesignal of the frequency k_(L)f_(x1) from the device 1A, a frequency ofthe local signal (LO signal) S5 from mpl2A which is given to RFMIX2 isset at (k_(L)+m) f_(x2) instead of k_(L)f_(x2). The reception signalthat is converted into the IF frequency has frequency converted by theIF frequency converter (IFMIX2), and an output signal S9 of a base bandis obtained. An output signal S4 of OSC2 is frequency-divided to asignal S6 having a frequency obtained by multiplying the output signalS4 by m by div2, and the signal S6 is used as an LO signal for IFMIX2. Aphase ϕ_(b2) of the signal S6 is expressed by equation (5) as follows.ϕ_(b2) =−m2πf _(x2) t+θ _(Bx2)  (5)Here, θ_(Bx2) is an initial phase of the LO signal for IFMIX2 from div2, and the frequency −mf_(x2) is an IF frequency.

In order to receive a signal from the device 1A in the device 2A, thephase ϕ_(tx2) of the output signal S5 of mpl2A is set at what is shownby equation (6) as follows that is obtained by transforming equation (4)described above.ϕ_(tx2)=2π(k _(L) +m)f _(x2) t+θ _(Lmx2(1))  (6)Here, θ_(Lmx2(1)) is an initial phase of the output signal S5 of mpl2Abetween the time t₁ and the time t₂. Note that it is not necessary tochange the frequency of mpl1A in the device 1A, and therefore the phaseϕ_(tx1) of the output signal S2 of mpl1A remains as in equation (2).

FIG. 6 is an explanatory diagram illustrating settings of the device 1Aand the device 2A between the time t₂ and the time t₃ in FIG. 3. Notethat in a period between the time t₂ and the time t₃, a receptionoperation of the device 2A is not performed, and therefore unnecessaryunits for operation are shown by dash-dotted lines.

In the device 1A that adopts a heterodyne method, RFMIX1 needs toconvert a reception signal into an IF frequency of approximately−mf_(x1). For this reason, in the device 1A that receives a single wavesignal of the frequency k_(L)f_(x2) from the device 2A, the frequency ofthe local signal (LO signal) S2 from mpl1A which is given to RFMIX1 isset at (k_(L)+m) f_(x1) instead of k_(L)f_(x1). The reception signalthat is converted into the IF frequency has frequency converted by theIF frequency converter (IFMIX1), and an output signal S12 is obtained.An output signal S1 of OSC1 is frequency-divided to a signal S3 having afrequency obtained by multiplying an output signal S1 of OSC1 by −m bydiv1, and the signal S3 is used as an LO signal for IFMIX1. A phaseϕ_(b1) of the signal S3 is expressed by equation (7) as follows.ϕ_(b1) =−m2πf _(x1) t+θ _(Bx1)  (7)Here, θ_(Bx1) is an initial phase of the LO signal for IFMIX1 from div1,and the frequency −mf_(x1) is an IF frequency.

In order to receive a signal from the device 2A, in the device 1A, thephase ϕ_(tx1) of the output signal S2 of mpl1A is set at what is shownby equation (8) as follows that is obtained by transforming equation (2)described above.ϕ_(tx1)=2π(k _(L) +m)f _(x1) t+θ _(Lmx1(1))  (8)Here, θ_(Lmx1(1)) is an initial phase of the output signal S2 of mpl1Abetween the time t₂ and the time t₃.

The device 2A returns the setting of the transmission frequency from(k_(L)+m) f_(x2) to k_(L)f_(x2) in a period between the time t₂ and thetime t₃. At this time, the phase ϕ_(tx2) of the output signal S5 ofmpl2A is expressed by equation (9) as follows. Note that θ_(Lx2(2)) isan initial phase of the signal S5 in this case.ϕ_(tx2)=2πk _(L) f _(x2t)+θ_(Lx2(2))  (9)

Settings of the device 1A and the device 2A are same as the settings inFIG. 5, between the time t₃ and the time t₄ that are in a next sequence.In the device 2A, in order to receive a single wave signal of thefrequency k_(L)f_(x1) from the device 1A, the frequency of the LO signal(signal SS) given to RFMIX2 is changed from k_(L)f_(x2) to (k_(L)+m)f_(x2). Note that in this case, the phase ϕ_(b2) of the LO signal(signal S6) for IFMIX2 given to IFMIX2 is the same as in equation (5)described above.

The phase ϕ_(tx2) of the output signal S5 of mpl2A of the device 2A isgiven by equation (10) as follows obtained by transforming equation (9)described above.ϕ_(tx2)=2π(k _(L) +m)f _(x2) t+θ _(Lmx2(2))  (10)Here, θ_(Lmx2(2)) is the initial phase of the output signal S5 of mpl2Abetween the time t₃ and the time t₄.

The device 1A returns the transmission frequency from (k_(L)+m) f_(x1)to k_(L)f_(x1). At this time, the phase ϕ_(tx1) of the output signal S2of mpl1A is set at what is shown by equation (11) as follows.ϕ_(tx1)=2πk _(L) f _(x1) t+θ _(Lx1(2))  (11)Here, θ_(Lx1(2)) is the initial phase of the output signal S2 of mpl1Abetween the time t₃ and the time t₄.

In this way, between the time t₁ and the time t₄, the phase ϕ_(tx1) ofthe signal S2 of mpl1A changes as shown by a thick line characteristicC1 in FIG. 4, and the phase ϕ_(tx2) of the signal S5 of mpl2A changes asshown by a thick line characteristic C2 in FIG. 4.

From a time D+t₁ to a time D+t₄ in FIG. 3, a single wave signal of thefrequency of k_(H)f_(x1) is transmitted from the device 1A, and a singlewave signal of the frequency of k_(H)f_(x2) is outputted from the device2A. Hereinafter, these frequencies are also referred to as highfrequencies. A sequence in this case only differs from the abovesequence in that k_(L) is changed to k_(H) in FIG. 5 and FIG. 6, andtherefore explanation will be omitted.

Next, referring to FIG. 5, a phase ϕ_(BB2L(T12)) (t) of the base bandsignal S9 detected in the device 2A between the time t₁ and the time t₂is obtained while phases of mpl1A and mpl2A are considered. A phaseϕ_(rx2) of a signal S7 after passing through the propagation path PDLYis expressed by equation (12) as follows.ϕ_(rx2)=2πk _(L) f _(x1)(t−τ_(R))+θ_(Lx1)  (12)Here, τ_(R) is a delay time of a propagation path length R. The signalS7 is frequency-converted by using the signal S5 (LO signal). Fromequation (12) and equation (6), a phase ϕ_(ifx2(T12)) (t) of an outputsignal S8 of RFMIX2 is expressed by equation (13) as follows.ϕ_(ifx2(T12))(t)=2πk _(L)(f _(x1) −f _(x2))t−2πmf _(x2)t+(θ_(Lx1)−θ_(Lmx2(1)))−2πk _(L) f _(x1)τ_(R)  (13)Note that equation (13) shows a phase result of extracting only adesired signal. The signal is frequency-converted by using the signalS6. Accordingly, from equation (13) and equation (5), the phaseϕ_(BB2L(T12)) (t) of the signal S9 detected in the device 2A is what isexpressed by equation (14) as follows.ϕ_(BB2L(T12))(t)=2πk _(L)(f _(x1) −f_(x2))t(θ_(Lx1)−θ_(Lmx2(1)))−θ_(Bx2)−2πk _(L) f _(x1)τ_(R)  (14)Note that equation (14) shows a result of performing desired quadraturedemodulation.

Similarly, with reference to FIG. 5, a phase of the signal S9 detectedin the device 2A between the time t₃ and the time t₄ is obtained. Fromequation (11), the phase of the signal S7 after passing through thepropagation path PDLY is expressed byϕ_(rx2)=2πk _(L) f _(x1)(t−τ _(R))+θ_(Lx1(2))  (15)from equation (11). The signal S7 is frequency-converted by the signalS5 (LO signal). From equation (15) and equation (10), a phaseϕ_(ifx2(T34)) (t) of an output signal S8 of RFMIX2 is expressed byequation (16) as follows.ϕ_(ifx2(T34))(t)=2πk _(L)(f _(x1) −f _(x2))t−2πmf _(x2)t+(θ_(Lx1(2))−θ_(Lmx2(2)))−2πk _(L) f _(x1)τ_(R)  (16)

Note that equation (16) shows a phase result of extracting only adesired signal. The signal is frequency-converted by using the signalS6. From equation (16) and equation (5), a phase ϕ_(BB2L(T34)) (t) ofthe signal S9 detected in the device 2A isϕ_(BB2L(T34))(t)=2πk _(L)(f _(x1) −f_(x2))t+(θ_(Lx1(2))−θ_(Lmx2(2)))−θ_(Bx2)−2πk _(L) f _(x1)τ_(R)  (17)Note that equation (17) describes a result of performing desiredquadrature demodulation.

Next, with reference to FIG. 6, a phase of the signal S12 detected inthe device 1A between the time t₂ and the time t₃ is obtained. A phaseϕ_(rx1) of a signal S10 after passing through the propagation path PDLYis given by equation (18) as follows from equation (9) described above.ϕ_(rx1)=2πk _(L) f _(x2)(t−τ _(R))+θ_(Lx2(2))  (18)

The signal S10 is frequency-converted by using the signal S2 (LO signal)in RFMIX1. From equation (18) and equation (8), a phase ϕ_(ifx1(T23))(t) of an output signal S11 of RFMIX1 is expressed by equation (19) asfollows.ϕ_(ifx1(T23))(t)=2πk _(L)(f _(x2) −f _(x1))t−2πmf _(x1)t+(θ_(Lx2(2))−θ_(Lmx1(1)))−2πk _(L) f _(x2)τ_(R)  (19)

Note that equation (19) shows a phase result of extracting only adesired signal. The signal S11 is frequency-converted by using thesignal S3. As a result, a phase ϕ_(BB2L(T23)) (t) of the signal S9detected in the device 2A is expressed by equation (20) as follows fromequation (19) and equation (7).ϕ_(BB2L(T23))(t)=2πk _(L)(f _(x2) −f_(x1))t+(θ_(Lx2(2))−θ_(Lmx1(1)))−θ_(Bx1)−2πk _(L) f _(x2)τ_(R)  (20)Note that equation (20) describes a result that a desired quadraturemodulation is performed.

Patent Literature 1 shows that a distance can be obtained by addition ofthe phases of the reception signals obtained by the distance measurementsequence. In the example of FIG. 3, an addition result ϕ_(BBLSUM) (t) offour phases is expressed by equation (21) as follows when the fourphases of reception signals obtained in transmission and reception ofsingle wave signals of a low frequency from the time t₁ to the time t₄are respectively ϕ_(12-1L), ϕ_(21-1L), ϕ_(21-2L), and ϕ_(12-2L.)ϕ_(BBLSUM)(t)=ϕ_(12-1L)+ϕ_(21-1L)+ϕ_(21-2L)+ϕ_(12-2L)  (21)

When an interval between the time t₂ and the time t₁ and an interval tobetween the time t₄ and the time t₃ are defined ast ₀ =t ₂ −t ₁ =t ₄ −t ₃  (22),and a time interval from a time at which a first distance measurementsignal is transmitted from the device 1A to a time at which a seconddistance measurement signal is transmitted from the device 2A is set asT, the four-phase addition result of equation (21) is as shown inequation (23) as follows.ϕ_(BBLSUM)(t)=ϕ_(BB2L(T12))(t)+ϕ_(BB2L(T23))(t+t₀)+ϕ_(BB2L(T23))(t+T)+ϕ_(BB2L(T34))(t+t ₀ +T)  (23)

Equation (14), equation (17) and equation (20) described above aresubstituted into equation (23) described above, and thereby equations(24) and (25) as follows are obtained.ϕ_(BBLSUM)(t)=4πk _(L)(f _(x1) +f_(x2))τ_(R)−2(θ_(Bx1)+θ_(Bx2))+θ_(LSUM)  (24)θ_(LSUM)=(θ_(Lx1)−θ_(Lmx2(1)))+2×(θ_(Lx2(2))−θ_(Lmx1(1)))+(θ_(Lx1(2))−θ_(Lmx2(2)))  (25)

When a delay τ_(R) is obtained from equation (24) described above, thedelay τ_(R) corresponding to a distance between devices is what is shownby equation (26) as follows.τ_(R)=(θ_(Bx1)+θ_(Bx2) 0/{2πk _(L)(f _(x1) +f _(x2))}−θ_(LSUM)/{4πk_(L)(f _(x1)+f_(x2))}+ϕ_(BBLSUM)(t)/{4πk _(L)(f _(x1) +f _(x2))}  (26)

A third term of equation (26) described above is the addition result ofthe four phases, and is obtained by measurement. However, the otherterms are difficult to detect. Accordingly, correct distance measurementcannot be performed with four alternations of single wave signals of alow frequency.

In the distance measurement sequence in FIG. 3, the devices 1A and 2Acarry out a sequence using single wave signals of a high frequencyfollowing the low frequency transmission. The high-frequency sequence issame as the low-frequency sequence, but a difference lies in changingthe frequency setting parameter k_(L) to k_(H). Important equations foranalysis will be shown as follows.

Between a time D+t₁ and a time D+t₂, the device 2A receives a singlewave signal of a frequency k_(H)f_(x1) from the device 1A. A phaseϕ_(BB2H(T12)) (t) of a signal S7 received by the device 2A is expressedby equation (27) as follows.ϕ_(BB2H(T12))(t)=2πk _(H)(f _(x1) −f_(x2))t(θ_(Hx1)−θ_(Hmx2(1)))−θ_(Bx2)−2πk _(H) f _(x1)τ_(R)  (27)Note that θ_(Hx1) is an initial phase of the signal S2 of the frequencyk_(H)f_(x1) of the device 1A, and θ_(Hmx2(1)) is an initial phase of thesignal S5 of a frequency (k_(H)+m)f_(x2) of the device 2A.

Between a time D+t₂ and a time D+t₃, the device 1A receives a singlewave signal of a frequency k_(H)f_(x2) from the device 2A. A phaseϕ_(BB2H(T23)) (t) of the signal S10 received by the device 1A isexpressed by equation (28) as follows.ϕ_(BB2H(T23))(t)=2πk _(H)(f _(x2) −f_(x1))t(θ_(Hx2(2()−θ_(Hmx1(1)))−θ_(Bx1)−2πk _(H) f _(x2)τ_(R)   (28)Note that θ_(Hx2(2)) is an initial phase of the signal S5 of thefrequency k_(H)f_(fx2) of the device 2A, and θ_(Hmx1(1)) is an initialphase of the signal S2 of the frequency (k_(H)+m)f_(x1) of the device1A.

Between a time D+t₃ and a time D+t₄, the device 2A receives a singlewave signal of a frequency k_(H)f_(x1) from the device 1A. A phaseϕ_(BB2H(T34)) (t) of the signal S7 received by the device 2A isexpressed by equation (29) as follows.ϕ_(BB2H(T34))(t)=2πk _(H)(f _(x1) −f_(x2))t+(θ_(Hx1(2))−θ_(Hmx2(2)))−θ_(Bx2)−2πk _(H) f _(x1)τ_(R)  (29)Note that an initial phase θ_(Hx1(2)) is an initial phase of the signalS2 of the frequency k_(H)f_(x1) of the device 1A, and θ_(Hmx2(2)) is theinitial phase of the signal S5 of the frequency (k_(H)+m)f_(x2) of thedevice 2A.

In the example of FIG. 3, an addition result ϕ_(BBHSUM) (t) of fourphases is expressed by equation (30) as follows when the four phases ofreception signals obtained in transmission and reception of single wavesignals of a high frequency from the time D+t₁ to the time D+t₄ arerespectively ϕ_(12-1H), ϕ_(21-1H), ϕ_(21-2H), and ϕ_(12-2H).ϕ_(BBHSUM)(t)=ϕ_(12-1H)+ϕ_(21-1H)+ϕ_(21-2H)+ϕ_(12-2H)  (30)

When equation (22) and the information on the time T are added toequation (30) described above, equation (31) as follows is obtained.ϕ_(BBHSUM)(t)=ϕ_(BB2H(T12))(t)+ϕ_(BB2H(T23))(t+t₀)+ϕ_(BB2H(T23))(t+T)+ϕ_(BB2H(T34))(t+T+t ₀)   (31)

When equation (31) is transformed by using equation (27), equation (28),and equation (29), equation (32) and equation (33) as follows areobtained.ϕ_(BBHSUM)(t)=−4πk _(H)(f _(x1) +f_(x2))τ_(R)−2(θ_(Bx1)+θ_(Bx2))+θ_(HSUM)  (32)θ_(HSUM)=(θ_(Hx1)−θ_(Hmx2(t)))+2×(θ_(Hx2(2))−θ_(Hmx1(1)))+(θ_(Hx1(2))−θ_(Hmx2(2)))  (33)

When the delay τ_(R) corresponding to the distance between the devicesis made a subject of equation (33), equation (34) as follows isobtained.τ_(R)=(θ_(Bx1)+θ_(Bx2))/{2πk _(H)(f _(x1) +f _(x2))}−θ_(HSUM)/{4πk_(H)(f _(x1) +f _(x2))}+ϕ_(BBHSUM)(t)/{4πk _(H)(f _(x1) +f _(x2))}  (34)

A third term of equation (34) is the addition result of the four phases,and can be detected by measurement. However, the other terms aredifficult to detect. Accordingly, correct distance measurement cannot beperformed with transmission and reception of four alternations by singlewave signals of a high frequency.

Next, distance measurement using two waves of a low frequency and a highfrequency is considered. In other words, the delay τ_(R) is obtained byperforming subtraction of equation (23) and equation (31) describedabove. Equation (35) as follows is obtained by subtraction of equation(23) and equation (31).ϕ_(BBLSUM)(t)−ϕ_(BBHSUM)(t)=4π(k _(H) −k _(L))(f _(x1) +f_(x2))τ_(R)+θ_(LSUM)−θ_(HSUM)  (35)

From equation (35), the delay τ_(R) is obtained by equation (36) asfollows.τ_(R)=−(θ_(LSUM)−θ_(HSUM))/4π(k _(H) −h _(L))(f _(x1) +f_(x2))+(ϕ_(BBLSUM)(t)−ϕ_(BBHSUM)(t))/4π(k _(H) −k _(L))(f _(x1) +f_(x2))  (36)

A second term of equation (36) is a value that is obtained by anoperation of the phases of the received single wave signals, that is, ameasurement value. However, a first term in equation (36) shows additionand subtraction of the initial phases of the signals S2 and S5 of thedevices 1A and 2A that are expressed by equation (25) and equation (33).The initial phases of the signals S2 and S5 are as shown in FIG. 4 inthe distance measurement sequence in FIG. 3. In the proposal of PatentLiterature 1, accurate distance measurement is possible by cancellingcomponents of the initial phase by using the condition that the initialphase does not change in the distance measurement sequence. However,when a DCO direct modulation method and a heterodyne method are used,the initial phase changes each time the frequency setting is changed asin FIG. 4, so that the first term of equation (36) described abovecannot be obtained, and the propagation delay time τ_(R) cannot beaccurately calculated. Since a distance can be calculated by multiplyingthe propagation delay time period by a light velocity, the distancecannot be accurately calculated in other words.

Note that the above described explanation shows the problem that thedistance measurement cannot be accurately performed due to thefluctuations of the initial phases of the output signals of mpl1A andmpl2A that are local oscillators in the distance measuring devices.However, it is conceivable that not only the distance measuring devicebut also various devices that detect the phases of signals by usinglocal oscillators may not be able to achieve desired functions due tofluctuation in the initial phases of the output signals. The presentembodiment is applicable to the various devices that detect the phasesof signals by using the local oscillators like this.

Correction Method of Initial Phase That Fluctuates

In the present embodiment, it is made possible to achieve a samefunction as in a case where an initial phase is not changed, in a deviceusing local oscillators, by adopting a reference phase device forobtaining a phase (hereinafter, referred to as a reference phase) thatchanges according to a frequency at an initial setting time from aninitial phase at a time of occurrence of a frequency of initial setting,that, is, an initial phase before performing resetting of the frequency,obtaining a fluctuation amount of the phase by an initial phase changeand a frequency change by obtaining a difference between the referencephase and the phase after resetting of the frequency, and correcting thephase according to the obtained fluctuation amount.

Distance Measuring Device

In FIG. 1, devices 1 and 2 that are distance measuring devices each havea configuration using a digital-controlled oscillator (DCO) directmodulation method for a transmission unit, and using a super heterodyne(SH) method for a reception unit. A distance measuring system 100 of thepresent embodiment includes the device 1 and the device 2, and at leastone of the device 1 and the device 2 is movable. The device 1 transmitsa first distance measurement signal (single wave signal), and the device2 transmits a second distance measurement signal (single wave signal).The first and the second distance measurement signals respectively reachthe device 2 and the device 1 via the propagation path PDLY between thedevice 1 and the device 2.

In FIG. 1, the device 1 has an oscillator (OSC1) peculiar to the device,a frequency multiplier (mpl1), an RF frequency converter (RFMIX1), afrequency divider (div1), and an intermediate (IF) frequency converter(IFMIX1). The device 2 has a same configuration as the configuration ofthe device 1, and has an oscillator (OSC2) peculiar to the device, afrequency multiplier (mpl2), an RF frequency converter (RFMDC2), afrequency divider (div2), and an intermediate (IF) frequency converter(IFMIX2).

In other words, a main point where the devices 1 and 2 respectivelydiffer from the devices 1A and 2A in FIG. 2 is that the devices 1 and 2respectively adopt mpl1 and mpl2 in place of mpl1A and mpl2A. In mpl1and mpl2, respective outputs are also used as local signals (LO signal).In other words, mpl1 and mpl2 respectively configure local oscillators.

An LO signal similar to the LO signal of mpl1A or mpl2A can be generatedby each of mpl1 and mpl2. Accordingly, in the present embodiment, thedistance measurement sequence illustrated in FIG. 3 can also be carriedout, and equation (36) described above obtaining the delay τ_(R)corresponding to the distance between the devices is established. Thepresent embodiment enables accurate distance measurement by obtainingthe value of the first term of equation (36) described above by adoptingmpl1 and mpl2.

First, with reference to a graph in FIG. 7, three kinds of phasedifferences that are necessary to calculate the first term of equation(36) described above will be described. FIG. 7 is a diagram in whicharrows explaining information on the five kinds of phase differences areadded to a graph similar to the graph in FIG. 4. Note that in FIG. 7 andFIG. 8 described later, TT included in subscripts in signs indicatingthe five kinds of phase differences indicates that a phase difference isrelated to a fluctuation in a phase of a signal, the frequency of whichis multiplied by k_(L), RR indicates that a phase difference is relatedto a fluctuation in a phase of a signal, the frequency of which ismultiplied by (k_(L)+m), and TR indicates that a phase difference isrelated to changes of the phase of the signal, the frequency of which ismultiplied by k_(L), and the phase of the signal, the frequency of whichis multiplied by (k_(L)+m). Further, L included in the subscripts in thesigns indicating the phase differences indicates that the single wavesignal has a low frequency, H indicates that a single wave signal has ahigh frequency, 1 indicates that a phase difference is related to thesignal S2, and 2 indicates that a phase difference is related to thesignal S5.

As described above, the devices 1 and 2 perform initial settings oftransmission frequencies by the time t₁ in FIG. 3. In other words, thedevice 1 and the device 2 respectively have the transmission frequenciesset at k_(L)f_(x1), and k_(L)f_(x2) by mpl1 and mpl2. Explaining thedistance measurement sequence in the low frequency in FIG. 3 again, theinitial phase of the output signal S2 of mpl1 of the device 1 changes toan initial phase θ_(Lx1) before the time t₂, an initial phaseθ_(Lmx1(1)) from the time t₂ to the time t₃, and an initial phaseθ_(Lx1(2)) from the time t₃ to the time t₄. As for the device 2, theinitial phase of the output signal S5 of mpl2 changes to an initialphase θ_(Lx2(2)) before the time t₁, an initial phase θ_(Lmx2(1)) fromthe time t₁ to the time t₂, an initial phase θ_(Lx2(2)) from the time t₂to the time t₃, and an initial phase θ_(Lmx2(2)) from the time t₃ to thetime t₄.

Phase differences Δθ_(LTT1) and Δθ_(LTR1) are phase differencesconcerning the device 1. The phase difference Δθ_(LTT1) is a differencebetween the initial phase θ_(Lx1(2)) from the time t₃ to the time t₄ andthe initial phase θ_(Lx1) before the time t₂ in the signal S2. The phasedifference Δθ_(LTR1) is a difference between the initial phaseθ_(Lmx1(1)) from the time t₂ to the time t₃ and the initial phaseθ_(Lx1) before the time t₂. Relationships among these variables can berespectively expressed by equation (37) and equation (38).θ_(Lx1(2))=θ_(Lx1)+Δθ_(LTT1)  (37)θ_(Lmx1(1))=θ_(Lx1)+Δθ_(LTR1)  (38)

Further, phase differences Δθ_(LTT2), Δθ_(LRR2), and Δθ_(LTR2) are phasedifferences concerning the device 2. The phase difference Δθ_(LTT2) is adifference between the initial phase θ_(Lx2(2)) from the time t₂ to thetime t₃ and the initial phase θ_(Lx2) before the time t₁ in the signalS5. The phase difference Δθ_(LRR2) is a difference between the initialphase θ_(Lmx2(2)) from the time t₃ to the time t₄ and the initial phaseθ_(Lmx2(1)) from the time t₁ to the time t₂ in the signal S5. The phasedifference Δθ_(LTR2) is a difference between the initial phaseθ_(Lmx2(1)) from the time t₁ to the time t₂ and the initial phaseθ_(Lx2) before the time t₁ in the signal S5. Relationships among thesevariables can be respectively expressed by equation (39) to equation(41) as follows.θ_(Lx2(2))=θ_(Lx2)+Δθ_(LTT2)  (39)θ_(Lmx1(1))=θ_(Lx2)+Δθ_(LTR2)  (40)θ_(Lmx2(2))=θ_(Lmx2(1))+Δθ_(LRR2)=θ_(Lx2)+Δθ_(LRT2)+Δθ_(LRR2)  (41)

As will be described later, of the phase differences, Δθ_(LTT1),Δθ_(LTT2), and Δθ_(LRR2) can be directly measured by mpl1 and mpl2. Onthe other hand, Δθ_(LTR1) in equation (38) and Δθ_(LTR2) in equation(40) cannot be directly measured. Therefore, in the present embodiment,mpl1 and mpl2 obtain Δθ_(LTR1) and Δθ_(LTR2) by measuring the phasedifferences relating to Δθ_(LTR1) and Δθ_(LTR2) as will be describedlater.

Here, in order to show a concept of an initial phase measurement method,Δθ_(LTR1) and Δθ_(LTR2) will be described as measurable.

When equation (37) to equation (41) described above are substituted intoθ_(LSUM) in equation (25) described above, θ_(LSUM) in equation (36)described above is given by equation (42) as follows.θ_(LSUM)=−2(Δθ_(LTR1)+Δθ_(LTR2))+2×Δθ_(LTT2)+Δθ_(LTT1)−Δθ_(LRR2)  (42)

Next, θ_(HSUM) in the high frequency shown in equation (33) is obtained.

A graph in FIG. 8 shows a transition of the initial phases of thesignals S2 and S5 in the distance measurement sequence of the highfrequency, and explains five kinds of phase differences that arenecessary for calculation of a first term in equation (36) describedabove, similarly to FIG. 7. FIG. 8 is a diagram in which arrowsexplaining information on the five kinds of phase differences are addedto a graph similar to the graph in FIG. 4. Note that the graph in FIG. 8illustrates an example having a characteristic of a same shape as theshape of the graph in FIG. 7 to simplify explanation, but does not haveto have the characteristic of the same shape as in FIG. 7.

The initial phase of the signal S2 from mpl1 of the device 1 changes toan initial phase θ_(Hx1) before a time t₂, an initial phase θ_(Hmx1(1))from the time t₂ to a time t₃, and an initial phase θ_(Hx1(2)) from thetime t₃ to a time t₄. The initial phase of the signal S5 from mpl2 ofthe device 2 changes to an initial phase θ_(Hx2) before a time t₁, aninitial phase θ_(Hmx2(1)) from the time t₁ to the time t₂, an initialphase θ_(Hx2(2)) from the time t₂ to the time t₃, and an initial phaseθ_(Hmx2(2)) from the time t₃ to the time t₄.

Phase differences Δθ_(HTT1) and Δθ_(HTR1) are phase differencesconcerning the device 1. The phase difference Δθ_(HTT1) is a differencebetween the initial phase θ_(Hx1(2)) from the time t₃ to the time t₄ andthe initial phase θ_(Hx1) before the time t₂ in the signal S2. The phasedifference Δθ_(HTR1) is a difference between the initial phaseθ_(Hmx1(1)) from the time t₂ to the time t₃ and the initial phaseθ_(Hx1) before the time t₂ in the signal S2. Relationships among thesevariables can be respectively expressed by equation (43) and equation(44) as follows.θ_(Hx1(2))=θ_(Hx1)+Δθ_(HTT1)  (43)θ_(Hmx1(1))=θ_(Hx1)+Δθ_(HTR1)  (44)

Likewise, phase differences Δθ_(HTT2), Δθ_(HRR2), and Δθ_(HTR2) arephase differences concerning the device 2. The phase differenceΔθ_(HTT2) is a difference between the initial phase θ_(Hx2(2)) from thetime t₂ to the time t₃ and the initial phase θ_(Hx2) before the time t₁in the signal S5. The phase difference Δθ_(HRR2) is a difference betweenthe initial phase θ_(Hmx2(2)) from the time t₃ to the time t₄ and theinitial phase θ_(Hmx2(1)) from the time t₁ to the time t₂ in the signalS5. The phase difference Δθ_(HTR2) is a difference between the initialphase θ_(Hmx2(1)) from the time t₁ to the time t₂ and the initial phaseθ_(Hx2) before the time t₁ in the signal S5. Relationships among thesevariables can be respectively expressed by equation (45) to equation(47) as follows.θ_(Hx2(2))=θ_(Hx2)+Δθ_(HTT2)  (45)θ_(Hmx2(1))=θ_(Hx2)+Δθ_(HTR2)  (46)θ_(Hmx2(2))=θ_(Hmx2(1))+Δθ_(HRR2)=θ_(Hx2)+Δθ_(HTR2)+Δθ_(HRR2)  (47)

As in the case of the low frequency, of the above phase differences,Δθ_(HTT1), Δθ_(HTT2), and Δθ_(HRR2) can be directly measured by mpl1 andmpl2. On the other hand, Δθ_(HTR1) in equation (44) and Δθ_(HTR2) inequation (46) cannot be directly measured. Therefore, in the presentembodiment, mpl1 and mpl2 obtain Δθ_(HTR1) and Δθ_(HTR2) by measuringthe phase differences relating to Δθ_(HTR1) and Δθ_(HTR2) as will bedescribed later.

Here, in order to show a concept of an initial phase measurement method,Δθ_(HRT1) and Δθ_(HTR2) will be described as measurable.

When equation (43) to equation (47) described above are substituted intoθ_(HSUM) in equation (25) described above, θ_(HSUM) in equation (36)described above is given by equation (48) as follows.θ_(HSUM)=−2(Δθ_(HTR1)+Δθ_(HRT2))+2×Δθ_(HTT2)+Δθ_(HTT1)−Δθ_(HRR2)  (48)

As above, it is possible to obtain the first term in equation (36)described above by equation (42) and equation (48) described above.

In thick line characteristics C1 to C4 in FIG. 7 and FIG. 8, sectionswhere inclinations are small are transmission sections, and sectionswhere inclinations are large are reception sections. Note that when oneof the devices 1 and 2 is in the transmission section, the other one isin the reception section. In equation (42) and equation (48) describedabove, the phase differences Δθ_(LTT1), Δθ_(LTT2), Δθ_(HTT1), andΔθ_(HTT2) express the phase differences between the respective RFsignals in the two transmission sections with the reception sectionsandwiched between the two transmission sections, in the respectivedevices 1 and 2. The phase differences Δθ_(LRR2), and Δθ_(HRR2) expressthe phase differences between the respective RF signals of the tworeception sections with the transmission section sandwiched between thetwo reception sections. The phase differences Δθ_(LTR1), Δθ_(LTR2),Δθ_(HTR1) and Δθ_(HTR2) each express the phase difference between therespective RF signals of the continuous transmission section andreception section. The former two are the initial phase differences ofthe same frequency, and the latter two are the initial phase differencesof different frequencies. When “the phase difference between therespective RF signals in the two transmission sections” (hereinafter,also referred to as a first phase difference), “the phase differencebetween the respective RF signals in the two reception sections”(hereinafter, also referred to as a second phase difference), and “thephase difference between the respective RF signals in the continuoustransmission section and reception section” (hereinafter, also referredto as a third phase difference) can be obtained, it becomes possible toperform accurate distance measurement by equation (36) described above.

In the present embodiment, the three kinds of phase differences orinformation for obtaining the three kinds of phase differences areobtained by mpl1 and mpl2. Information concerning the obtained phasedifferences is outputted to the operation devices CA1 and CA2 by mpl1and mpl2 respectively. The operation devices CA1 and CA2 arerespectively given signals S12 and S9 from IFMIX1 and IFMIX2, and detectphases of the signals S12 and S9. The operation device CA1 performs anoperation of equation (36) described above to obtain the delay τ_(R) andfurther obtains the distance R, by using phase information obtained fromthe signal S12 and information concerning the phase difference frommpl1. Note that in the device 2, the operation device CA2 can alsoperform an operation of equation (36) described above to obtain thedelay τ_(R) and further obtain the distance R, by using phaseinformation obtained from the signal S9 and the information concerningthe phase difference from mpl2. Note that the operation devices CA1 andCA2 can respectively perform various kinds of control concerningdistance measurement in the devices 1 and 2, for example, frequencysetting, timing control and the like in the distance measurementsequence.

Specific Configuration

FIG. 9 is a block diagram illustrating the distance measuring deviceincluding the phase fluctuation detecting device and the phasecorrecting device according to the embodiment, and illustrates aspecific configuration of mpl2 that calculates the informationconcerning the above described three kinds of phase differences in thedevice 2. Further, the configuration of mpl1 of the device 1 is alsosimilar to the configuration in FIG. 9, and illustration and explanationwill be omitted. Note that in FIG. 9, the phase fluctuation detectingdevice is mainly configured by OSC2 and mpl2, and the phase correctingdevice is configured by OSC2, mpl2, a phase calculator phscalc2, and adistance calculator dcalc2. As described above, it is possible to usethe phase fluctuation detecting device and the phase correcting devicenot only in the distance measuring device, but also in various devicesthat detect phases of inputted signals, and in that case, in the phasecorrecting device, other circuits that correct the phase of the inputsignal according to the fluctuation amount of the initial phase by usingthe output of mpl2 are adopted, instead of the phase calculator phscalc2and the distance calculator dcalc2.

A frequency multiplier mpl20, a reference phase device mpl00, and aphase detector phsdet configure mpl2. The frequency multiplier mpl20 hasa same function as the function of mpl2A in FIG. 2. In other words, thefrequency multiplier mpl20 configures a local oscillator, is given asignal S4 that is an oscillation output of OSC2, multiplies a frequencyof the signal S4 by a predetermined amount, generates and outputs thesignal S5 that is a local oscillation signal. Note that a multiplicationnumber by the frequency multiplier mpl20 is designated by a controldevice CN2 of the operation device CA2. The control device CN2 generatesfrequency control data for determining the multiplication number of thefrequency multiplier mpl20.

The signal S5 is given to RFMIX2 as an LO signal in the receptionsection of the distance measurement, and is transmitted as the singlewave signal in the transmission section of the distance measurement. Thefrequency multiplier mpl20 can also output information on a phase of thesignal S5 to the phase detector phsdet.

The reference phase device mpl00 is given the frequency control datafrom the control device CN2. As described above, in the distancemeasurement sequence, a frequency of the signal S5 from the frequencymultiplier mpl20 changes, and an initial phase of the signal S5 alsochanges at a timing of a change of the frequency. In the presentembodiment, the reference phase device mpl00 is given frequency controldata (hereinafter, referred to as reference frequency control data)before an oscillation frequency of the frequency multiplier mpl20 ischanged. Thereby, the reference phase device mpl00 can outputinformation on a phase that changes according to the initial phase andan initial frequency before the frequency of the signal S5 from thefrequency multiplier mpl20 is changed, that is, a phase for obtainingthe reference phase (hereinafter, referred to as a quasi-referencephase). The reference phase device mpl00 outputs the obtainedquasi-reference phase to the phase detector phsdet.

The phase detector phsdet acquires information for obtaining the abovedescribed three kinds of phase differences based on the inputtedinformation, and outputs the information (S15) to the operation deviceCA2.

The operation device CA2 is configured by a phase calculator phscalc2, adistance calculator dcalc2 and a control device CN2. The control deviceCN2 controls operations of the phase calculator phscalc2 and thedistance calculator dcalc2 that configure a correction circuit, andcontrols mpl2 and div2. The control device CN2 is capable of frequencycontrol, timing control and the like concerning distance measurement inthe device 2, and can also set the aforementioned frequency controldata, for example.

The phase calculator phscalc2 obtains θ_(LSUM) and θ_(HSUM) of equation(36) described above to output θ_(LSUM) and θ_(HSUM) to the distancecalculator dcalc2, by using the output of the phase detector phsdet. Theoperation device CA2 is also given a signal S9 from IFMIX2, and thedistance calculator dcalc2 obtains the delay τ_(R) by an operation ofequation (36) described above from the output of the phase calculatorphscalc2 and the signal S9, and further calculates the distance R.

FIG. 10 is a circuit diagram illustrating a more specific configurationof part of mpl2 and the phase calculator phscalc2. Note that aconfiguration of part of mpl1 and the phase calculator phscalc1 of thedevice 1 is also similar to the configuration in FIG. 10, andillustration and explanation will be omitted.

A frequency multiplier mpl20 includes a circuit part of a frequencymultiplier of an ordinary configuration including an ADPLL (all digitalphase-locked loop) including a digitally controlled oscillator (DCO).The digitally controlled oscillator DCO generates an oscillation outputof an oscillation frequency corresponding to an inputted digital valueand outputs the oscillation output. As will be described later, at atime of lock of the ADPLL, the digitally controlled oscillator DCOgenerates an oscillation output of a frequency that is a rationalmultiple of a frequency of the reference clock that is generated by thereference oscillator 10. Note that the reference oscillator 10corresponds to OSC2 in FIG. 1 and FIG. 9.

The oscillation output of the digitally controlled oscillator DCO issupplied to a counter 11. The counter 11 counts the oscillation outputof the digitally controlled oscillator DCO, and a count value of thecounter 11 is outputted to a subtractor 12. The counter 11 counts anumber of waves (number of pulses) of the oscillation output of thedigitally controlled oscillator DCO. A count value of the counter 11 inone period of the reference clock indicates how many integer multiplesof the reference clock, for example, the oscillation output of thedigitally controlled oscillator DCO is.

The oscillation output of the digitally controlled oscillator DCO isalso supplied to TDC13. TDC13 may be configured by a plurality of delayelements of a delay time sufficiently shorter than the period of theoscillation output. TDC13 is also given the reference clock, and TDC13obtains a delay time (corresponding to a phase difference) between theoscillation output of the digitally controlled oscillator DCO and thereference clock, and outputs the delay time to a normalization circuit14. The normalization circuit 14 normalizes the output of TDC13 with oneperiod of the reference clock as 1. In other words, an output of thenormalization circuit 14 indicates that how many decimal multiples ofthe reference clock period the output (delay time) of TDC13 is, andindicates the phase difference between the output of the digitallycontrolled oscillator DCO and the reference clock. The output of thenormalization circuit 14 is supplied to the subtractor 12.

An integrator (Σ) 15 is given frequency control data and the referenceclock. The frequency control data indicates a multiplication number of arational number to the reference clock, which is a value of a ratio of adesired oscillation output frequency of the digitally controlledoscillator DCO and a reference clock frequency. The integrator 15integrates the frequency control data at each reference clock, andoutputs an integration result to the subtractor 12.

An output of the counter 11 is an integration result of an integermultiplication number of the frequency of the output of the digitallycontrolled oscillator DCO to the reference clock, and the output of thenormalization circuit 14 is a decimal multiplication number of thefrequency of the output of the digitally controlled oscillator DCO tothe reference clock. The outputs of the counter 11 and the normalizationcircuit 14 each indicate a multiplication number of a rational number ofthe frequency of the output of the digitally controlled oscillator DCOthat is oscillating to the reference clock. The outputs of the counter11 and the normalization circuit 14 each indicate a present phase of theoutput of the digitally controlled oscillator DCO with the referenceclock as a reference.

The subtractor 12 obtains a phase error by subtracting the outputs ofthe counter 11 and the normalization circuit 14 from an output of theintegrator 15. The subtractor 12 gives the obtained phase error to thedigitally controlled oscillator DCO via a loop filter 16 and anormalization circuit 17. Thereby, the oscillation output of thedigitally controlled oscillator DCO changes in frequency so that anoutput of the subtractor 12 becomes zero. Note that the loop filter 16operates at reference clock periods, and the normalization circuit 17normalizes an output of the loop filter 16 to information suitable forfrequency control of the digitally controlled oscillator DCO and givesthe information to the digitally controlled oscillator DCO. In this way,at a time of lock of the ADPLL, an oscillation output of a frequency ofa rational number multiple based on the frequency control data of thereference clock is obtained from the digitally controlled oscillatorDCO.

As described above, the outputs of the counter 11 and the normalizationcircuit 14 each indicate the present phase of the output of thedigitally controlled oscillator DCO with the reference clock as areference, the output of the counter 11 indicates a present phase of aninteger multiple of 2π (360 degrees), and the output of thenormalization circuit 14 indicating a decimal multiplication numberindicates a present phase at a time of the output of the digitallycontrolled oscillator DCO with the reference clock as the referencebeing normalized by 2π, that is, setting 2π as 1. At the time of lock,the output of the subtractor 12 becomes zero, so that the output of theintegrator 15 also indicates a present phase of the output of thedigitally controlled oscillator DCO with the reference clock as thereference.

The output of the integrator 15 is also outputted to a subtractor 30.

An integrator (Σ) 20 configures the reference phase device mpl00 in FIG.9. The integrator 20 is given reference frequency control data. Theintegrator 20 is also given the reference clock from the referenceoscillator 10. The integrator 20 integrates the reference frequencycontrol data at each reference clock, and outputs an integration resultto the subtractor 30, similarly to the integrator 15. The referencefrequency control data is data of an initial value before the frequencycontrol data that is supplied to the integrator 15 is changed.

In other words, the integrator 20 can output information on a phase thatchanges according to an initial phase and an initial frequency beforethe frequency of the signal S5 from the frequency multiplier mpl20 beingchanged, that is, a phase for obtaining a reference phase (hereinafter,referred to as a quasi-reference phase).

The phase detector phsdet in FIG. 9 is configured by the subtractor 30,and a part of the phase calculator phscalc2 in FIG. 9 is configured by atiming generation circuit 40, hold circuits 44 and 45, a subtractor 46,and a MOD (remainder operation) circuit 47. The subtractor 30 subtractsan output of the integrator 20 from the output of the integrator 15, andoutputs a subtraction result to the hold circuits 44 and 45. When aphase indicated by the output of the integrator 15 (hereinafter,referred to as an output phase of the integrator 15) is set as ϕ₂, and aphase indicated by the output of the integrator 20 (hereinafter,referred to as an output phase of the integrator 20) is set as ϕ₀. thesubtractor 30 obtains a difference between ϕ₂ and ϕ₀. Note that in thepresent description, ϕ₂ and ϕ₀ are treated as signals obtained bynormalizing 2π to 1 on the circuit.

The output phase (quasi-reference phase) of the integrator 20 is a phasethat changes at a similar change rate to a change rate of the referencephase and matches in initial phase or differs only in initial phase withor from the phase (reference phase) of the output of the frequencymultiplier mpl20 at the time of the initial setting, that is, the outputphase of the integrator 15. Accordingly, when the frequency control datathat is inputted to the integrator 15 is not changed from an initialvalue, the frequency control data and the reference frequency controldata have same values as each other, and an output of the subtractor 30becomes 0 or a predetermined fixed value and does not change.

For example, it is assumed that when the frequency of the referenceoscillator 10 is f_(x2) and the frequency control data of the initialvalue is K_(a), the output phase θ (t) of the integrator 15 is expressedby θ (t)=K_(a)f_(x2)t+θ_(a1). θ_(a1) is an initial phase in this case.In this case, the output phase θ₀ (t) of the integrator 20 is expressedby θ₀ (t)=K_(a)f_(x2)t+θ₀. θ₀ is an initial phase in this case. In thiscase, the output of the subtractor 30 is θ_(a1)−θ₀, and is a fixedvalue.

When the frequency control data that is supplied to the integrator 15changes to K_(b), the output phase θ (t) of the integrator 15 becomes θ(t)=K_(b)f_(x2)t+θ_(b). Note that θ_(b) is an initial phase in thiscase, and the output of the subtractor 30 changes toK_(b)f_(x2)t+θ_(b)−(K_(a)f_(x2)t+θ₀). When the frequency control datareturns to K_(a), the output phase θ (t) of the integrator 15 becomes θ(t)=K_(a)f_(x2)t+θ_(a2). Note that θ_(a2) is an initial phase in thiscase, and the output of the subtractor 30 changes to θ_(a2)−θ₀.

In this way, the output of the subtractor 30 corresponds to the changein the frequency and the initial phase, and it is possible to remove aninfluence by the initial phase by using the output of the subtractor 30.When applied to the distance measuring device, the output of thesubtractor 30 is supplied to the hold circuits 44 and 45 that configurea part of the phase calculator phscalc2, as information S15 foracquiring the aforementioned first to third phase differences.

The output of the integrator 15 corresponds to the output phase of thedigitally controlled oscillator DCO, and corresponds to the phaseϕ_(tx2) shown by the thick line characteristics C2 and C4 in FIG. 7 andFIG. 8. When illustrated with θ₀=θ_(a1) for easy understanding of therelationship, the output phase (quasi-reference phase) of the integrator20 corresponds to a straight line of the phase ϕ_(tx2) shown by a dashedline with θ_(Lx2) or θ_(Hx2) in FIG. 7 and FIG. 8 as an initial phase.The output of the subtractor 30 is a difference between the thick linecharacteristic C2 in FIG. 7 and FIG. 8 and the quasi-reference phaseexpressed by a straight line.

The timing generation circuit 40 is given the reference clock andgenerates predetermined timing signals ta1 and ta2 to output the timingsignals ta1 and ta2 to the hold circuits 44 and 45, with the referenceclock as a reference.

FIG. 11 is a block diagram illustrating one example of a specificconfiguration of the timing generation circuit 40. The timing generationcircuit 40 is configured by a counter 41, a decoder 42 and a delaydevice 43. The counter 41 counts the reference clock to output a countvalue to the decoder 42. The decoder 42 generates two timing signalscorresponding to the count value to output the two timing signals to thedelay device 43 by being controlled by the control device CN2 of theoperation device CA2 (not illustrated), or based on information storedin a memory not illustrated. The delay device 43 generates the timingsignals ta1 and ta2 by delaying the inputted two timing signals by apredetermined delay time. Note that the timing signals ta1 and ta2 willbe described later.

The hold circuit 44 outputs a phase θ_(A) acquired by holding an outputof the subtractor 30 at timing of the timing signal ta1 to thesubtractor 46. The hold circuit 45 outputs a phase θ_(B) acquired byholding an output of the subtractor 30 at timing of the timing signalta2 to the subtractor 46. Note that the timing generation circuit 40sets times at which the output frequency of the frequency multipliermpl20 is stabilized in the reception section and the transmissionsection as the timings ta1 and ta2.

The subtractor 46 performs subtraction of the phases θ_(A) and θ_(B),and outputs a subtraction result to the MOD circuit 47. As describedlater, the aforementioned first to third phase differences can beobtained from the subtraction result of the subtractor 46. For example,it is obvious that Δθ_(LTT2) in FIG. 7 is easily obtained based on anoutput of the subtractor 46.

FIG. 7 and FIG. 8 show that the phase ϕ_(tx2) shown by the thick linecharacteristic C2 and the quasi-reference phase simply increase, but inreality, the phase does not exceed 2π. The MOD circuit 47 obtains aremainder by 2π of the output of the subtractor 46, and outputs Δθ_(AB)that is information on a phase difference. A distance measurementoperation is performed by using Δθ_(AB).

When 2π is treated as 1 as in the present embodiment, the MOD circuit 47can be a circuit that takes out only a decimal portion from an inputtedsignal. Alternatively, the integrator 20, the subtractor 30, the holdcircuit 44, the hold circuit 45, the subtractor 46 and the like may bemade the circuits that handle only decimal portions, without the MODcircuit 47 being provided, and it is obvious that a circuit scale can bereduced by doing so.

Next, an operation of the embodiment that is configured in this way willbe described with reference to a graph in FIG. 12. FIG. 12 is a graph bya similar description method to the description method in FIG. 7, inwhich the characteristics concerning the device 1 (device 1A) is removedfrom the graph in FIG. 7, and shows an output phase (quasi-referencephase) of the integrator 20 that configures the reference phase devicempl00 by ϕ₀. It is assumed that mpl2 operates similarly to mpl2A of thedevice 2A. In other words, the phase of the output of mpl2 of the device2, that is, the output phase ϕ₂ of the integrator 15 is shown by FIG. 12(characteristic C2) showing a similar characteristic to thecharacteristic of FIG. 7. Note that the integrator 15 and the integrator20 operate discretely at each period of a reverse number of a frequencyf_(x2) of the reference oscillator 10, and therefore, strictly speaking,output phases of the integrators change stepwise. The graph in FIG. 12has a staircase shape when it is enlarged enough, but in this case, thegraph is expressed such that stairs are omitted for the sake ofintuitive understanding. The same applies to the following graphs. Inthe present description, to make the explanation easier to understand,in the following equations and explanations, explanation is performedfor a behavior at each time at which f_(x2)t or f_(x1)t becomes aninteger, that is, at each discrete time will be described.

It is assumed that the reference phase device mpl00 and mpl20 have thesame frequency at the time of the initial setting of the transmissionfrequency before the time t₁, and a similar initial setting to theinitial setting of mpl2A described above is performed. Accordingly, theoutput phase ϕ₂ of the integrator 15 indicating a phase ϕ_(tx2) of theoutput of mpl20 is equivalent to a right side of equation (4) describedabove, and is expressed by a thick line characteristic C2 in FIG. 12. Inmpl20, k_(L) in equation (4) means a rational number indicated byfrequency control data K_(a). An integer value (integer multiplicationnumber) of the rational number corresponds to a multiple of 360° (2π) inphase conversion, and is omitted in a remainder operation for obtainingthe phase difference Δθ_(AB). Therefore, in using the output phase ϕ₂ ofthe integrator 15 corresponding to the phase ϕ_(tx2) of the output ofmpl20, it is not necessary to consider the integer multiplicationnumber, and only a change in the phase by a decimal value (decimalmultiplication number) may be used, though it is a repetition of theabove. Note that in the following explanation, the right side ofequation (4) including a phase amount by the integer multiplicationnumber is directly used, but there is no particular problem.

Calculation of First Phase Difference

The phase ϕ₂ of the output of the integrator 15 is given by equation(49) as follows that is similar to the right side of equation (4).ϕ₂=2πk _(L) f _(x2) t+θ _(Lx2)  (49)

In the reference phase device mpl00, a frequency setting in the initialsetting is same as the frequency setting in the initial setting ofmpl20, but an initial phase does not have to be same as the initialphase of mpl20. When the initial phase at the time of a low frequency ofthe reference phase device mpl00 is θ_(L0x2), a quasi-reference phase ϕ₀of the output of the integrator 20 that is obtained from the output ofthe reference phase device mpl00 is given by equation (50) as follows.ϕ₀=2πk _(L) f _(x2) t+θ _(L0x2)  (50)

Handling of the integer multiplication number is similar to the case ofϕ₂, and the integer multiplication number is assumed to be included inequation (50). When ϕ₂−ϕ₀ is detected in the subtractor 30 immediatelybefore the time t₁ at which the frequency is switched,ϕ₂−ϕ₀=θ_(Lx2)−θ_(L0x2)  (51)is established, and a difference between the initial phase of thefrequency multiplier mpl20 and the initial phase of the reference phasedevice mpl00 in the initial setting is obtained.

As described above, in the device 2, a period from the time t₁ to thetime t₂ is the reception section, and there is the transmission sectionagain after the time t₂, so that an output frequency of the frequencymultiplier mpl20 changes, and the initial phase of the output of thedigitally controlled oscillator DCO also changes. The phase ϕ₂ of theoutput of the integrator 15 from the time t₂ to a time t₃ is given byequation (52) as follows that is similar to a right side of equation (9)described above.ϕ₂=2πk _(L) f _(x2) t+θ _(Lx2(2))  (52)

The quasi-reference phase ϕ₀ of the output of the integrator 20 that isthe output of the reference phase device mpl00 follows equation (50)described above, because the frequency setting is not changed in thereference phase device mpl00. Accordingly, when ϕ₂−ϕ₀ is detected by thesubtractor 30 at a time at which the frequency is stabilized in thetransmission section from the time t₂ to the time t₃, equation (53) asfollows is obtained.ϕ₂−ϕ₀=θ_(Lx2(2))−θ_(L0x2)  (53)

A difference between ϕ₂−ϕ₀ that is detected immediately before the timet₁ and ϕ₂−ϕ₀ that is detected at the time when the frequency isstabilized in the transmission section from the time t₂ to the time t₃shows a phase difference Δθ_(LTT2) that is a change in the initial phasedue to a change in the output frequency of the frequency multipliermpl20. In other words, equation (51) is subtracted from equation (53),whereby θ_(Lx2(2))−θ_(Lx2) is obtained, and this is Δθ_(LTT2).

Processing of the above will be described with a circuit operation. Now,the phases of the frequency multiplier mpl20 and the reference phasedevice mpl00 that are detected at a time t_(A) immediately before thetime t₁ are respectively set as ϕ₂ (t_(A)), and ϕ₀ (t_(A)), and thephases of the frequency multiplier mpl20 and the reference phase devicempl00 that are detected at a time T_(B) between the time t₂ and the timet₃ are respectively set as ϕ₂ (t_(B)), and ϕ₀ (t_(B)).

The timing generation circuit 40 sets the time t_(A) immediately beforethe time t₁ as the timing ta1, and sets the time t_(B) between the timet₂ and the time t₃ as the timing ta2. Thereby, the hold circuit 45outputs a phase ϕ_(B)=ϕ₂ (t_(B))−ϕ₀ (t_(B)), and the hold circuit 44outputs θ_(A)=ϕ₂ (t_(A))−ϕ₀ (t_(A)).

In this way, the subtractor 46 obtains θ_(B)−θ_(A). In other words, theoutput of the subtractor 46 is Δθ_(LTT2) that is expressed by equation(54) as follows.Phase difference Δθ_(LTT2)=ϕ₂(t _(B))−ϕ₀(t _(A))−{ϕ₂(t _(A))−ϕ₀(t_(A))}  (54)

In this way, Δθ_(LTT2) that is a change in the initial phase of thefrequency multiplier mpl20, that is, the first phase difference (“thephase difference between the respective RF signals in the twotransmission sections”) is outputted from the subtractor 46. For theoutput of the subtractor 46, a remainder of 2π is obtained by the MODcircuit 47. Thereby the MOD circuit 47 outputs Δθ_(LTT2) as Δθ_(AB).

Note that from a MOD circuit 47 of the device 1 not illustrated,Δθ_(LTT1) is obtained by a method similar to the above described method.

Calculation of Second Phase Difference

Next, with reference to FIG. 13, FIG. 14 and FIG. 15, a method fordetecting the “the phase difference between the respective RF signals inthe two reception sections” that is the second phase difference will bedescribed. FIG. 13 is a similar graph to the graph in FIG. 12. In otherwords, in FIG. 13, the characteristic concerning the device 1 (device1A) is removed from the graph in FIG. 7, and FIG. 13 shows the outputphase (quasi-reference phase) ϕ₀ of the integrator 20 that is the outputof the reference phase device mpl00. It is assumed that mpl2 operatessimilarly to mpl2A of the device 2A. In other words, a phase of theoutput of mpl2 of the device 2, that is, the output phase ϕ₂ of theintegrator 15 is shown by a characteristic C2 in FIG. 13.

The phase ϕ_(tx2)(t) of an LO signal from the frequency multiplier mpl20in the reception section of the device 2 from the time t₃ to the time t₄is obtained from the output of the integrator 15. The output phase ϕ₂(t)of the integrator 15 in this case is expressed by equation (55) asfollows that is similar to a right side of equation (10) describedabove.ϕ₂(t)=ϕ_(tx2)(t)=2π(k _(L) +m)f _(x2) t+ϕ _(Lmx2(2))  (55)

Here, θ_(Lmx2(2)) is an initial phase of the output signal S5 of mpl20in a section from a time t₃ to a time t₄. An initial phase of the outputsignal S5 of mpl20 in a section from a time t₁ to a time t₂ is set asθ_(Lmx2(1)), a phase jump amount at a time of the device 2 transitionsfrom reception to transmission is set as Δθ_(JP2), and a phase jumpamount at a time of the device 2 transitioning from transmission toreception is set as Δθ_(JP3). If there is no change in the frequency, atotal of the phase jump amounts Δθ_(JP2) and Δθ_(JP3) is equal to adifference between θ_(Lmx2(2)) and θ_(Lmx2(1)), and when the frequencychange is taken into consideration, a relationship between θ_(Lmx2(2))and θ_(Lmx2(1)) is expressed by equation (56) as follows.θ_(Lmx2(2))−θ_(Lmx2(1)) =−mf _(x2)×(t ₃ −t ₂)+Δθ_(JP2)+Δθ_(JP3)  (56)

FIG. 14 is an explanatory diagram for explaining a difference between aphase ϕ_(tx2) of mpl20 and a quasi-reference phase ϕ₀ from the referencephase device mpl00.

FIG. 14 shows a value obtained by performing remainder calculation of 2πfor a phase difference between the phase ϕ_(tx2) (ϕ₂) and thequasi-reference phase ϕ₀ in a vertical axis. Note that the remainder isapplied to simplify explanation of a phase difference detection methodof a reception state. To simplify the explanation,Δθ_(JP2)=Δθ_(JP3)=0[rad] is set.

In other words, a waveform in FIG. 14 shows a change in ϕ₂ in a case ofthe quasi-reference phase ϕ₀ being set as a reference. When a phase in atransmission state (initial setting state) before a time t1 is set asϕ_(Lt1)(t), a phase in a reception section from the time t₁ to a time t₂is set as ϕ_(Lt1-t2)(t), a phase in a transmission section from the timet₂ to a time t₃ is set as ϕ_(Lt2-t3)(t), and a phase in a receptionsection from the time t₃ to a time t₄ is set as ϕ_(Lt3-t4)(t), waveformsshowing the phase differences in the respective sections in FIG. 14 canbe respectively expressed by equation (57) to equation (60) as follows,as seen from FIG. 13. Note that an IF frequency f_(IF2) of the device 2is a frequency difference mf_(x2)[Hz] between an LO signal at atransmission time and an LO signal at a reception time, and equation(61) as follows is established.ϕ_(Lt1)(t)=mod(θ_(Lx2)−θ_(LOx2), 2π)=θ_(TXA)  (57)ϕ_(Lt1-t2)(t)=mod(2πmf _(x2) t+θ _(Lmx2(1))−θ_(LOx2), 2π)=mod(2πf _(IF2)t+θ _(Lmx2(1))−θ_(LOx2), 2π)  (58)ϕ_(Lt2-t3)(t)=mod(θ_(Lx2(2))−θ_(LOx2), 2π)=θ_(TXB)  (59)ϕ_(Lt3-t4)(t)=mod(2πmf _(x2) t+θ _(Lmx2(2))−θ_(LOx2), 2π)=mod(2πf _(IF2)t+θ _(Lmx2(2))−θ_(LOx2), 2π)  (60)f_(IF2)=mf_(x2)  (61)

As shown in FIG. 14, a phase difference between the phase of the outputof mpl20 and the quasi-reference phase is constant in the transmissionsections, and changes at a change rate of an IF angular frequency(2πmf_(x2)) in the reception sections. Here, a case where the receptionstate does not end at the time t₂, and reception is continued until thetime t₄ is considered. In this case, equation (58) described above isalso applied to a section from the time t₂ to the time t₄.

FIG. 15 is an explanatory diagram illustrating waveforms of phasedifferences to which equation (58) is applied by adding the waveforms bya dash-dotted line to the waveforms of the phase differences in FIG. 14.A time period (one period) T_(IF2) in which the phase changes from0[rad] to 2π[rad] is 1/IF frequency, that is,T _(IF2)=1/f _(IF2)  (62).When an arbitrary time when the frequency is stabilized in the receptionsection from a time t₁ to a time t₂ is set as t_(IFA), if reception ofthe signal of the frequency f_(IF2) is continued thereafter, the phaseof the output of mpl20 has a same value at each time period T_(IF2), asshown by the dash-dotted line in FIG. 15. In other words, a phasedifference between the phase of the output of mpl20, the frequency ofwhich is assumed not to change and the quasi-reference phase has a samevalue at each time period T_(IF2) from the time t_(IFA).

Accordingly, when a difference between the output phase ϕ₂ of theintegrator 15 that indicates the output phase of mpl20 and thequasi-reference phase ϕ₀ is obtained in the subtractor 30, if a timet_(IFB) is assumed to be a time after an integer multiple of T_(IF2)from the time t_(IFA), in the section from a time t₃ to a time t₄ inFIG. 15, and a phase difference between the output phase of mpl20detected at the time t_(IFB) and the quasi-reference phase is set asΔθ(t_(IFB))v (dashed line), a phase difference Δθ(t_(IFA)) and the phasedifference Δθ(t_(IFB))v (dashed line) have a same value.

However, since the frequency changes in the reception section from atime t₂ to the time t₃ in reality, the phase of the output of mpl20changes according to equation (60), and the phase difference Δθ(t_(IFB))at the time t_(IFB) has a different value from the value of the phasedifference Δθ(t_(IFA)).

The difference of the phase differences in the case of receiving thefrequency change and the case of not receiving the frequency change inthe section from the time t₂ to the time t₃ is due to the fact that theinitial phase changes with the frequency change in the section from thetime t₂ to the time t₃ because there is no change in the frequency inthe two reception sections. In other words, an initial phase fluctuationamount Δθ_(LRR2)=Δθ(t_(IFB))−Δθ(t_(IFB))v is established. In theoperation, the quasi-reference phase is cancelled out, and therefore,the initial phase fluctuation amount Δθ_(LRR2) is obtained by equation(63) as follows using ϕ_(Lt2-t4)(t_(IFB)) that is the output phase ϕ₂ ofthe integrator 15 at the time t_(IFB), and ϕ_(Lt1-t2)(t_(IFA)) that isthe output phase ϕ₂ of the integrator 15 at the time t_(IFA).Δθ_(LRR2)=ϕ_(Lt3-t4)(t_(IFB))−ϕ_(Lt1-t2)(t_(IFA))  (63)

Note that equation (63) described above is obtained by setting both thephase jumps Δθ_(JP2), and Δθ_(JP3) as 0[rad]. However, as is obviousfrom FIG. 13, the phase ϕ_(Lt3-t4)(t_(IFB)) reflects Δθ_(JP2)+Δθ_(JP3),and even when the phase jumps Δθ_(JP2), and Δθ_(JP3) are not 0[rad], aninitial phase fluctuation amount Δθ_(LRR2) is obtained by equation (63)described above.

In other words, in the present embodiment, the timing generation circuit40 sets the time t_(IFA) as the timing signal ta1, and sets the timet_(IFB) as the timing signal ta2. The hold circuit 44 outputs 42 ϕ₂−ϕ₀at the time t_(IFA) as θ_(A), and the hold circuit 45 outputs ϕ₂−ϕ₀ atthe time t_(IFB) as θ_(B).

The subtractor 46 calculates the fluctuation amount Δθ_(LRR2) of theinitial phase based on θ_(B)−θ_(A). The MOD circuit 47 takes a remainderby 2π of the output of the subtractor 46, and outputs the fluctuationamount Δθ_(LRR2) as Δθ_(AB). In this way, it is possible to detect “thephase difference between the respective RF signals of the two receptionsections” that is the second phase difference.

Note that the initial phase fluctuation amount Δθ_(LRR2) is alsoobtained by an operation of a difference between the output of theintegrator 15 at the time T_(IFA) in the first reception section, andthe output of the integrator 15 at the time t_(IFB) in the secondreception section after a time period that is an integer multiple ofT_(IF2) from the time t_(IFA).

Note that from the MOD circuit 47 of the device 1 not illustrated,Δθ_(LRR1) is obtained by a similar method to the method described above.

Calculation of Third Phase Difference

Next, with reference to FIG. 16, a method for detecting the “phasedifference between the respective RF signals in the transmission sectionand the reception section that are continuous” that is the third phasedifference will be described. FIG. 16 is a similar explanatory diagramto the diagram in FIG. 14. Note that the third phase difference is phasedifferences Δθ_(LTR1), Δθ_(LTR2), Δθ_(HTR1), and Δθ_(HTR2) in FIG. 7 andFIG. 8. Of the phase differences, the phase difference Δθ_(LTR2)concerning transition from the section before the time t₁ of the device2 to the reception section from the time t₁ to the time t₂ will bedescribed as an example hereinafter, but as for the other cases, thephase differences can be similarly obtained.

FIG. 16 is a diagram illustrating a phase difference between thequasi-reference phase ϕ₀ and the output phase ϕ₂ of the integrator 15indicating the output phase of the frequency multiplier mpl20 in thedevice 2 by applying a remainder of 2π by a similar method to the methodin FIG. 14. Hereinafter, explanation will be performed by setting samplepoints at a time t_(C) before the time t₁ and a time t_(D) after thetime t₁.

A phase difference detected at the time t_(C) is mod(θ_(Lx2)−θ_(LOx2),2π) as shown in equation (57) described above. A phase differencedetected at the time t_(D) isϕ_(Lt1-t2)(t _(D))=mod(2πf _(IF2) t _(D)+θ_(Lmx2(1))−θ_(LOx2), 2π)  (64)from equation (58). When the equation is simplified to omit mod, thephase detected at the time t_(C) is set as ϕ_(Lt1)(t_(C)), and adifference between the two phase differences given from equation (58)and equation (64) is taken, equation (65) as follows is obtained.ϕ_(Lt1-t2)(t _(D))−ϕ_(Lt1)(t _(C))=2πf _(IF2) t_(D)+θ_(Lmx2(1))−θ_(Lx2)  (65)

From equation (40) and equation (65) described above, equation (66) asfollows is obtained.Δθ_(LTR2)=θ_(Lmx2(1))−θ_(Lx2)=ϕ_(Lx2)=ϕ_(Lt1-t2)(t _(D))−ϕ_(Lt1)(t_(C))−2πf _(IF2) t _(D)  (66)

From the output of the subtractor 30, ϕ_(Lt1-t2)(t_(D)) andϕ_(Lt2)(t_(C)) in equation (66) described above are obtained.Accordingly, when the IF frequency f_(IF2) and the time t_(D) aredefined, Δθ_(LTR2) can be obtained from equation (66) described above.

Note that equation (66) described above is obtained with a phase jump ata time of transition from a section before the time t₁ to a section fromthe times t₁ to t₂ set as 0 [rad]. However, as is obvious from FIG. 13,the phase ϕ_(Lt1-t2)(t_(D)) at the time t_(D) reflects the phase jump,and even when the phase jump is not 0 [rad], the initial phasefluctuation amount Δθ_(LTR2) is obtained by equation (66) describedabove.

In other words, in the present embodiment, the IF frequency f_(IF2) andthe time t_(D) in the first reception section are defined, and thetiming generation circuit 40 sets the time t_(C) in the output sectionby the initial setting as the timing signal ta1, and sets the time t_(D)as the timing signal ta2. The hold circuit 44 obtains ϕ₂−ϕ₀ at the timet_(C) as the phase θ_(A), and the hold circuit 45 obtains ϕ₂−ϕ₀ at thetime t_(D) as the phase θ_(B).

The subtractor 46 obtains the initial phase fluctuation amount Δθ_(LTR2)based on θ_(B)−θ_(A). The MOD circuit 47 obtains the remainder by 2π ofthe output of the subtractor 46, and outputs the fluctuation amountΔθ_(LTR2) of the initial phase as a phase difference Δθ_(AB).

Note that by an operation including a difference between ϕ₂−ϕ₀ at thetime t_(C) and ϕ₂−ϕ₀ at the time t_(D), the initial phase fluctuationamount Δθ_(LTR2) can also be detected.

From the MOD circuit 47 of the device 1 not illustrated, Δθ_(LTR1) isobtained by a similar method to the method described above.

Next, Δθ_(HTR2) is obtained. The distance measurement sequence of thelow frequency and the distance measurement sequence of the highfrequency in FIG. 3 only differ in frequency and start time of thesequence, and the time sequences in transmission and reception are same.It is assumed that the reference phase device mpl00 and mpl20 have asame frequency at the times of the initial settings after the time t₄and before D+t₁ as in the content described in [0099]. In other words,when a difference in frequency is ignored, the phase difference betweenthe quasi-reference phase of the device 2 and the output phase of theintegrator 15 indicating the phase of the output of the frequencymultiplier mpl20 can be expressed by a similar waveform to the waveformin FIG. 16 in the distance measurement sequence of the high frequency,and with respect to a sequence start time t=0 [s] of the low frequency,the start time can be offset by t=D [s] in the sequence of the highfrequency.

Accordingly, equations in which L expressing the meaning of the lowfrequency is changed to H expressing the meaning of the high frequency,and the detection time of the phase is changed from the time t_(D) tothe time D+t_(D) and is changed from the time t_(C) to the time D+t_(C)in equation (64) to equation (66) described above are established.

In other words, Δθ_(HTR2) can be expressed by equation (67) as followsthat is obtained by transforming equation (66) described above bysetting the phase of the output of the frequency multiplier mpl20 at thetime D+t_(D) as ϕ_(Ht1-t2)(D+t_(D)) and by setting the phase of theoutput of the frequency multiplier mpl20 at the time D+t_(C) as ϕ_(Ht1)(D+t_(C)).Δθ_(HTR2)=θ_(Hmx2(1))−θ_(Hx2)=ϕ_(Ht1-t2)(D+t _(D))−ϕ_(Ht1)(D+t _(C))−2πf_(IF2) t _(D)  (67)

Note that the IF frequency of the device 2 hardly changes whether the IFfrequency is at a high frequency or at a low frequency, final terms ofequation (66) and equation (67) have a same value.

In equation (67) described above, ϕ_(Hft1-t2)(D+t_(D)) and ϕ_(Ht1)(D+t_(C)) in equation (67) described above are obtained by the outputsof the subtractor 30. Accordingly, when the IF frequency f_(IF2) and thetime t_(D) are defined, Δθ_(HTR2) can be obtained from equation (66)described above. Equation (67) is established regardless of a size of aphase jump at a time of transition from the section before the time t₁to the section from the times D+t₁ to D+t₂.

In other words, in the present embodiment, the IF frequency f_(IF2) andthe time t_(D) in the first reception section are defined, and thetiming generation circuit 40 sets the time D+t_(C) in the output sectionby the initial setting as the timing signal ta1, and sets the timeD+t_(D) as the timing signal ta2. The hold circuit 44 obtains ϕ₂−ϕ₀ atthe time D+t_(C) as the phase θ_(A), and the hold circuit 45 obtainsϕ₂−ϕ₀ at the time D+t_(D) as the phase θ_(B). The subtractor 46 obtainsthe initial phase fluctuation amount Δθ_(HTR2) by θ_(B)−θ_(A). The MODcircuit 47 obtains the remainder by 2π of the output of the subtractor46, and outputs the initial phase fluctuation amount Δθ_(HTR1) as thephase difference Δθ_(AB).

Note that it is also possible to calculate the fluctuation amountΔθ_(HTR1) based on a difference between an output of the integrator 15at the time D+t_(C), and an output of the integrator 15 at the timeD+t_(D).

From the MOD circuit 47 of the device 1 not illustrated, Δθ_(HTR1) isobtained by a similar method to the method described above.

In this way, it is possible to detect the “phase difference between therespective RF signals in the transmission section and the receptionsection that are continuous” that is the third phase difference. Thephase calculator phscalc2 and the distance calculator dcalc2 of thedevice 2 performs a distance measurement operation by using the obtainedfirst to third phase differences, that is, the initial phase fluctuationamounts.

Distance Measurement Calculation

The θ_(LSUM) in equation (36) described above can be calculated by usingthe first to the third phase differences as shown in equation (42)described above. Likewise, the θ_(HSUM) in equation (36) described abovecan also be calculated by using the first to the third phase differencesas shown in equation (48) described above. The phase calculator phscalc2outputs θ_(LSUM) and θ_(HSUM) that are calculated to the distancecalculator dcalc2. The distance calculator dcalc2 obtains the delayτ_(R) by the operation of equation (36) described above from the outputof the phase calculator phscalc2 and the signal S9, and furthercalculates a distance R.

Although the subtractor 30 is described as performing the operation ofthe IF frequency f_(IF2) and the time t_(D) at the time of calculationof the above described third phase difference, the operation can beomitted as will be shown as follows. Equation (68) as follows expressesθ_(LSUM)−θ_(HSUM) in the first term of equation (36) described above,from equation (42) and equation (48).θ_(LSUM)−θ_(HSUM)=−2(Δθ_(LTR1)−Δθ_(HTR1))−2(Δθ_(LTR2)−Δθ_(HTR2))+2(Δθ_(LTT2)−Δθ_(HTT2))+(Δθ_(LTT1)−Δθ_(HTT1))−(Δθ_(LRR2)−Δθ_(HRR2))  (68)

A first term and a second term of equation (68) each show “the phasedifference between the respective RF signals in the transmission sectionand the reception section that are continuous” that is the third phasedifference, a third term and a fourth term each show “the phasedifference between the respective RF signals in the two transmissionsections” that is the first phase difference, and a fifth term shows“the phase difference between the respective RF signals in the tworeception sections” that is the second phase difference. In other words,the second term is a difference between “the phase difference betweenthe respective RF signals in the transmission section and the receptionsection that are continuous” using the low frequency and “the phasedifference between the respective RF signals in the transmission sectionand the reception section that are continuous” using the high frequency,in the device 2. When a difference between equation (66) and equation(67) is taken in order to obtain the difference, equation (69) asfollows is obtained.Δθ_(LTR2)−Δθ_(HTR2)=ϕ_(Lt1-t2)(t _(D))−ϕ_(Lt1)(t _(C))−{ϕ_(Ht1-t2)(D+t_(D))−ϕ_(Ht1)(D+t _(C))}  (69)

In equation (69), the terms of the operation of the IF frequency f_(IF2)and the time t_(D) are cancelled out. In other words, it shows that whenthe predetermined time t_(D) with the frequency setting start as thereference is set at a fixed value, the difference in the “phasedifference of the respective RF signals in the transmission section andthe reception section that are continuous” that is the third phasedifference can be obtained by the subtractor 30 only obtaining thedifference of the output phases of the integrators 15 and 20.

In the device 1, it is also possible to obtain Δθ_(LTR1)−Δθ_(HRT1) ofthe first term by a similar method.

Accordingly, in this case, the phase calculator phscalc2 can calculateθ_(LSUM)−θ_(HSUM) by using an operation result of the subtractor 30, andoutput a calculation result to the distance calculator dcalc2.

Note that as is obvious from FIG. 16, in the case where the frequency isnot stabilized at the time t_(D), the phase can be sampled at a timet_(D)+T_(IF2) that is delayed by the IF period.

In the explanation so far, calculation of the first phase difference,calculation of the second phase difference, and calculation of the thirdphase difference are described with separate timings. It is necessary toobtain ϕ_(Lt2-t3) (t)−ϕ_(Lt1) (t) in the calculation of the first phasedifference, ϕ_(Lt3-t4) (t_(IFB))−ϕ_(Lt1-t2) (t_(IFA)) in the calculationof the second phase difference, and ϕ_(Lt1-t2) (t_(D))−ϕ_(Lt1) (t_(C))or the like in the calculation of the third phase difference. Sincethese calculations are in parallel with one another in terms of time,the phase calculator phscalc2 can obtain the respective values byincluding a plurality of sets of the timing generation circuits 40, thehold circuits 44 and 45, the subtractors 46 and the MOD circuits 47.

Here, the case of performing the above described circuit by one set asin FIG. 10 is considered. it is assumed that the hold circuit 44 holdsϕ_(Lt1) (t_(C)), the hold circuit 45 holds a value three times, and thesubtractor 46 and the MOD circuit 47 output Δθ_(AB) that is a result ofsubtracting the value of ϕ_(Lt1) (t_(C)) held by the hold circuit 44from the subtractor 46 and the MOD circuit 47 to the distance calculatordcalc2 three times. When phases at the three times of holding are set asϕ_(Lt1-t2) (t_(IFA)), ϕ_(Lt2-t3) (t), and ϕ_(Lt3-t4) (t_(IFB)) in a timesequence, and Δθ_(AB) at the respective phases are set as Δθ_(AB1),Δθ_(AB2), and Δθ_(AB3), Δθ_(AB1), Δθ_(AB2), and Δθ_(AB3) can beexpressed by equations (70) to (72). Note that the equations areexpressed by being simplified and having mod omitted.Δθ_(AB1)=ϕ_(Lt1-t2)(t _(IFA))−ϕ_(Lt1)(T _(C))  (70)Δθ_(AB2)=ϕ_(Lt2-t3)(t)−ϕ_(Lt1)(t _(C))  (71)Δθ_(AB3)=ϕ_(Lt3-t4)(t _(IFB))−ϕ_(Lt1)(t _(C))  (72)When Δθ_(AB3)−Δθ_(AB1) is calculated here, equation (73) is obtained.Δθ_(AB3)−Δθ_(AB1)=ϕ_(Lt3-t4)(t _(IFB))−ϕ_(Lt1-t2)(t _(IFA))  (73)

When the equations are looked at here, it is found that equation (71)corresponds to the calculation of the first phase difference, equation(73) corresponds to the calculation of the second phase difference, andequation (70) corresponds to the calculation of the third phasedifference in a case of t_(D)=T_(IFA). Therefore, it is found that ifequation (73) is calculated by the distance calculator dcalc2, anecessary phase difference can be obtained with the configuration inFIG. 10, and the distance can be calculated.

In this way, in the present embodiment, it is possible to achieve thesimilar function to the function in the case of not changing the initialphase, in the device that detects the phase of a signal by using thelocal oscillator by obtaining the fluctuation amount of the phase due tothe initial phase change and the frequency change by adopting thereference phase device for obtaining the reference phase and obtainingthe difference between the quasi-reference phase obtained from theoutput of the reference phase device and the phase of the output afterresetting of the frequency, and correcting the phase according to theobtained fluctuation amount.

For example, when the present embodiment is applied to a distancemeasuring device that performs transmission and reception of single wavesignals between devices and performs distance measurement from areception phase, and is a distance measuring device using a directmodulation method for a transmission unit and using a super heterodynemethod for a reception unit, a fluctuation amount of an initial phasefollowing a frequency change in a distance measurement sequence can bedetected and corrected, and therefore accurate distance measurement ispossible from phase information.

Note that the present invention is not limited to the above describedembodiment, and can be modified variously in the range without departingfrom the gist of the present invention in the implementation stage. Theabove described embodiment includes the inventions in various stages,and various inventions can be extracted by appropriate combinations inthe plurality of components that are disclosed. For example, even whensome components are deleted from all the components shown in theembodiment, the configuration from which the components are deleted canbe extracted as the invention when the problem described in the columnof the problem to be solved by the invention can be solved, and theeffect described in the column of the effect of the invention isobtained.

While certain embodiments have been described, these embodiments havebeen presented by way of example only, and are not intended to limit thescope of the inventions. Indeed, the novel devices and methods describedherein may be embodied in a variety of other forms; furthermore, variousomissions, substitutions and changes in the form of the embodimentsdescribed herein may be made without departing from the spirit of theinventions. The accompanying claims and their equivalents are intendedto cover such forms or modification as would fall within the scope andspirit of the inventions.

What is claimed is:
 1. A phase correcting device, comprising: a localoscillator that includes an all digital phase-locked loop configured togenerate a local oscillation signal based on a reference clock, and isconfigured to give the local oscillation signal to a device configuredto detect a phase of an inputted signal; a first phase detector includedin the all digital phase-locked loop, and configured to detect a phaseof the local oscillation signal to output the phase of the localoscillation signal; a reference phase device configured to generate aquasi-reference phase corresponding to a reference phase of the localoscillation signal at a time of an initial setting of the localoscillator to output the quasi-reference phase, based on the referenceclock; a second phase detector configured to detect a fluctuation amountof a phase of the local oscillator, based on the phase detected by thefirst phase detector and the quasi-reference phase; and a correctioncircuit configured to correct the phase of the inputted signal by usinga detection result of the second phase detector.
 2. The phase correctingdevice according to claim 1, wherein the first phase detector includes afirst integrator configured to be given frequency control data fordesignating a multiplication number of the local oscillation signal, andthe reference phase device includes a second integrator configured to begiven frequency control data for designating a multiplication number ofthe local oscillation signal at the time of the initial setting of thelocal oscillator.
 3. The phase correcting device according to claim 2,wherein the second phase detector includes a first subtractor configuredto obtain a difference between an output of the first integrator and anoutput of the second integrator.
 4. The phase correcting deviceaccording to claim 3, wherein the second phase detector comprises holdcircuits configured to hold outputs of the first subtractor atpredetermined two timings different from each other, and a secondsubtractor configured to obtain a difference between the outputs of thefirst subtractor that are held at the two timings.
 5. The phasecorrecting device according to claim 1, wherein the all digitalphase-locked loop comprises the first phase detector, a digital controloscillator configured to control a frequency by digital control based onfrequency control data, a counter circuit and a time-digital conversioncircuit configured to obtain a phase of the local oscillation signalfrom the digital control oscillator with a phase of the reference clockas a reference, and a subtractor configured to obtain a difference ofoutputs of the first phase detector that is given the frequency controldata, and the counter circuit and the time-digital conversion circuit tocontrol the digital control oscillator based on a difference result, andthe first phase detector gives an output at a time of lock of the alldigital phase-locked loop to the second phase detector as a detectionresult of the phase of the local oscillation signal.
 6. A distancemeasuring device that calculates a distance based on carrier phasedetection, comprising: an operation device configured to calculate adistance between a first device and a second device based on phaseinformation acquired from the first device and the second device, atleast one of the first device and the second device being movable,wherein the first device comprises a first local oscillator thatincludes a first all digital phase-locked loop configured to generate afirst local oscillation signal based on a first reference clock, and isconfigured to output the first local oscillation signal, a firsttransmitter configured to transmit two or more first carrier signals byusing an output of the first local oscillator by a direct modulationmethod, a first receiver configured to receive two or more secondcarrier signals by using an output of the first local oscillator by aheterodyne method, a first output phase detector included in the firstall digital phase-locked loop, and configured to detect a phase of thefirst local oscillation signal to output the phase of the first localoscillation signal, a first reference phase device configured togenerate a first quasi-reference phase corresponding to a firstreference phase of the first local oscillation signal at a time of aninitial setting of the first local oscillator to output the firstquasi-reference phase, based on the first reference clock, and a firstfluctuation phase detector configured to detect a first fluctuationamount of a phase of the first local oscillator, based on a phasedetected by the first output phase detector and the firstquasi-reference phase, the second device comprises a second localoscillator that includes a second all digital phase-locked loopconfigured to generate a second local oscillation signal based on asecond reference clock, and is configured to output the second localoscillation signal, a second transmitter configured to transmit the twoor more second carrier signals by using an output of the second localoscillator by a direct modulation method; a second receiver configuredto receive the two or more first carrier signals by using an output ofthe second local oscillator by a heterodyne method, a second outputphase detector included in the second all digital phase-locked loop, andconfigured to detect a phase of the second local oscillation signal tooutput the phase of the second local oscillation signal, a secondreference phase device configured to generate a second quasi-referencephase corresponding to a second reference phase of the second localoscillation signal at a time of an initial setting of the second localoscillator to output the second quasi-reference phase, based on thesecond reference clock, and a second fluctuation phase detectorconfigured to detect a second fluctuation amount of a phase of thesecond local oscillator, based on a phase detected by the second outputphase detector, and the second quasi-reference phase, and the operationdevice performs calculation of the distance based on a phase detectionresult obtained by reception of the first carrier signal and the secondcarrier signal by the first receiver and the second receiver, and thefirst fluctuation amount and the second fluctuation amount detected bythe first fluctuation phase detector and the second fluctuation phasedetector.
 7. A phase fluctuation detecting device, comprising: a localoscillator that includes an all digital phase-locked loop configured togenerate a local oscillation signal based on a reference clock, and isconfigured to give the local oscillation signal to a device configuredto detect a phase of an inputted signal; a first phase detector includedin the all digital phase-locked loop, and configured to detect a phaseof the local oscillation signal to output the phase of the localoscillation signal; a reference phase device configured to generate aquasi-reference phase corresponding to a reference phase of the localoscillation signal at a time of an initial setting of the localoscillator to output the quasi-reference phase, based on the referenceclock; and a second phase detector configured to detect a fluctuationamount of a phase of the local oscillator, based on a phase detected bythe first phase detector and the quasi-reference phase.
 8. A phasecorrection method, comprising: giving a local oscillation signal to adevice configured to detect a phase of an inputted signal, from a localoscillator including an all digital phase-locked loop configured togenerate the local oscillation signal based on a reference clock;detecting a phase of the local oscillation signal to output the phase ofthe local oscillation signal, by a first phase detector included in theall digital phase-locked loop; generating a quasi-reference phasecorresponding to a reference phase of the local oscillation signal at atime of an initial setting of the local oscillator to output thequasi-reference phase, based on the reference clock, by a referencephase device, detecting a fluctuation amount of a phase of the localoscillator based on a phase detected by the first phase detector and thequasi-reference phase by a second phase detector, and correcting thephase of the inputted signal by using a detection result of the secondphase detector.